Cancellation of burst noise in a communication system with application to S-CDMA

ABSTRACT

A communication system performs burst noise cancellation. A transmitter produces and transmits a spread signal that comprises at least one known-value symbol spread by a plurality of non data-carrying orthogonal codes and data symbols spread by at least one data-carrying orthogonal code. The transmitter transmits the spread signal across a communication link that introduces burst noise. A burst noise detector determines burst noise affected chips of the orthogonal codes. A weight computation functional block calculates a plurality of complex-valued combining weights based upon the burst noise affected chips. A vector de-spreader and a linear combiner operate in combination to use the plurality of non data-carrying orthogonal codes, the at least one data-carrying orthogonal code, and the plurality of complex-valued combining weights to de-spread the received spread signal to produce the data symbols with the burst noise substantially removed.

CROSS REFERENCE TO RELATED PATENTS/PATENT APPLICATIONS ContinuationPriority Claim, 35 U.S.C. § 120

The present U.S. Utility patent application claims priority pursuant to35 U.S.C. § 120, as a continuation, to the following U.S. Utility patentapplication which is hereby incorporated herein by reference in itsentirety and made part of the present U.S. Utility patent applicationfor all purposes:

1. U.S. Utility application Ser. No. 11/089,139, entitled “Cancellationof burst noise in a communication system with application to S-CDMA,”filed Mar. 24, 2005, now issued as U.S. Pat. No. 7,415,061 B2 on Aug. 192008, which claims priority pursuant to 35 U.S.C. § 119(e) to thefollowing U.S. Provisional Patent Application which is herebyincorporated herein by reference in its entirety and made part of thepresent U.S. Utility patent application for all purposes:

1.a. U.S. Provisional Application Ser. No. 60/647,182, entitled“Cancellation of burst noise in a communication system with applicationto S-CDMA,”, filed Jan. 26, 2005, now expired, and the U.S. Utilityapplication Ser. No. 11/089,139 also claims priority pursuant to 35U.S.C. § 120, as a continuation-in-part (CIP), to the following U.S.Utility patent application which is hereby incorporated herein byreference in its entirety and made part of the present U.S. Utilitypatent application for all purposes:

2. U.S. Utility application Ser. No. 10/142,189, entitled “Cancellationof interference in a communication system with application to S-CDMA,”,filed May 8, 2002, now U.S. Pat. No. 7,110,434, issued on Sep. 19, 2006,which claims priority pursuant to 35 U.S.C. § 119(e) to the followingU.S. Provisional Patent Application which is hereby incorporated hereinby reference in its entirety and made part of the present U.S. Utilitypatent application for all purposes:

2.a. U.S. Provisional Application Ser. No. 60/367,564, entitled“Cancellation of interference in a communication system with applicationto S-CDMA,” filed Mar. 26, 2002, now expired, and the U.S. Utilityapplication Ser. No. 10/142,189 also claims priority pursuant to 35U.S.C. § 120, as a continuation-in-part (CIP), to the following U.S.Utility patent application which is hereby incorporated herein byreference in its entirety and made part of the present U.S. Utilitypatent application for all purposes:

3. U.S. Utility application Ser. No. 09/652,721, entitled“Subdimensional single-carrier modulation,” filed Aug. 31, 2000, nowU.S. Pat. No. 6,778,611, issued on Aug. 17, 2004, which claims prioritypursuant to 35 U.S.C. § 119(e) to the following U.S. Provisional PatentApplication which is hereby incorporated herein by reference in itsentirety and made part of the present U.S. Utility patent applicationfor all purposes:

3.a. U.S. Provisional Application Ser. No. 60/151,680, entitled“Subdimensional single-carrier modulation,” filed Aug. 31, 1999, nowexpired.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

The invention relates generally to communication systems; and, moreparticularly, it relates to communication systems that may be affectedby undesirable interference.

2. Description of Related Art

Signal processing within communication systems having a communicationchannel, in an effort to improve the quality of signals passing throughthe communication channel, has been under development for many years. Inthe past several years, emphasis has moved largely to the domain ofdigital communication systems that modulate bit streams into an analogsignal for transmission over a communication channel. This channel canbe a variety of channel types. Many different approaches are employed inthe prior art to try to minimize or substantially reduce the effects ofinterference that may be introduced into a signal that is transmittedacross a communication channel. In particular, the prior art approachesthat seek to perform cancellation of interference that occupies a smallnumber of signal dimensions in a signal are typically deficient for anumber of reasons as is briefly referenced below. One particular type ofinterference that these prior art schemes seek to minimize is thenarrowband interference that is sometimes referred to as ingressinterference. Another type of interference occupying only a subset ofthe dimensions used by the signaling waveform is the interference ofimpulse/burst noise. Yet another type of interference that may beproblematic is within the code division multiple access (CDMA) contextwhen the interference is on a small number of codes. In the presentcontext, the term “code,” “spreading code” or “despreading code” refersto a sequence of chips which are used to spread or despread a signal,such as in a spread-spectrum or CDMA system. This usage should not beconfused with the language used to describe forward error correction(FEC) coding.

One of the main methods employed in the prior art to eliminatenarrowband interference is the use of a notch filter. This solution issufficient in some applications, but the notch filter itself oftentimescauses distortion of the desired signal. In the CDMA context, thisdistortion is called inter-code interference (ICI). Then, another meansmust oftentimes be included to remove the very ICI that has beenintroduced by the notch filter. One way to do this is to de-spread thesignal. Then, hard decisions are made using the de-spread signal. Thehard decisions are then respread and passed through the notch filter andsubtracted from the original signal to remove the estimated distortion.In some instances, this process is repeated numerous times to try toachieve an adequate result. These prior art approaches described aboveare deficient in that they suffer the effect of error propagation. Thedecision circuit is prone to make incorrect decisions, requiring manyiterations before the process converges, if it ever converges at all.

Similar problems exist for iterative methods to remove impulse or burstnoise. The excision of the chips associated with the impulse or burstnoise may be thought of as a time-domain filter. This filter itselfcauses distortion of the desired signal. In the CDMA context, thisdistortion is called inter-code interference (ICI). Then, another meansmust oftentimes be included to remove the very ICI that has beenintroduced by the filter. One way to do this is to de-spread the signal.Then, hard decisions are made using the de-spread signal. The harddecisions are then respread and passed through the filter and subtractedfrom the original signal to remove the estimated distortion. In someinstances, this process is repeated numerous times to try to achieve anadequate result. These prior art approaches described above aredeficient in that they suffer the effect of error propagation. Thedecision circuit is prone to make incorrect decisions, requiring manyiterations before the process converges, if it ever converges at all.

Various techniques have been used to overcome problems caused by burstnoise interference. These techniques include Forward Error Correction(FEC) coding, Automatic Retransmission reQuest (ARQ) operations, andtime spreading. FEC coding includes adding coded (or parity) bits in thetransmitter. At the receiver, the parity bits are used to reconstructthe missing pieces of the signal that resulted from degradation by burstnoise. An example of FEC coding is block coding such as Reed-Solomoncoding, in which an N-symbol block is sent consisting of k informationsymbols and N−k parity symbols. A burst of noise corrupting up toT=(N−k)/2 Reed-Solomon symbols can be corrected, and if erasure decodingis used, up to 2T Reed-Solomon symbols can be marked for erasure and theinformation is still recovered. With ARQ operations, error detection atthe receiver determines missing data packets (or portions thereof) thatwere affected by the burst noise and causes retransmission of suchmissing data packets. ARQ operations introduce latency due to therequirements of lost data detection and retransmission.

Time spreading in Code Division Multiple Access (CDMA) systems, e.g., inan S-CDMA DOCSIS 2.0 system, spreads each data symbol across a timeperiod. A given noise burst therefore may appear only during a partialduration of the data symbol. The effect of the noise burst can berepaired by coding and/or other techniques. Time spreading has a limitedrange over which it operates effectively, since burst noise that is verystrong, that is of long duration compared to a spread symbol (e.g., manychips), or that corrupts more FEC symbols than the FEC is capable ofcorrecting, can overwhelm the communication system.

Further limitations and disadvantages of conventional and traditionalsystems will become apparent through comparison of such systems with theinvention as set forth in the remainder of the present application withreference to the drawings.

BRIEF SUMMARY OF THE INVENTION

A communication system that is operable to perform burst noisecancellation helps to solve the above-described problems among others.The burst noise cancellation may operate alone, or in conjunction withthe above-mentioned elements such as FEC and iterative processing. Thecommunication system includes a transmitter and a receiver. Thetransmitter is operable to produce a spread signal that comprises atleast one known-value symbol (which in one embodiment may be zero)spread by a plurality of non data-carrying orthogonal codes and datasymbols spread by at least one data-carrying orthogonal code. Thetransmitter is further operable to transmit the spread signal across acommunication link. The receiver is operable to receive the spreadsignal after being transmitted across the communication link. Uponreceipt by the receiver, the received spread signal may include burstnoise. The receiver includes a burst noise detector, a weightcomputation functional block, a vector de-spreader, and a linearcombiner. The burst noise detector is operable to determineburst-noise-affected chips of the orthogonal codes. The weightcomputation functional block is operable to calculate a plurality ofcomplex-valued combining weights based upon the burst-noise-affectedchips. The vector de-spreader and the linear combiner are operable incombination to use the plurality of non data-carrying orthogonal codes,the at least one data-carrying orthogonal code, and the plurality ofcomplex-valued combining weights to de-spread the received spread signalto produce the data symbols with the burst noise substantially removed.Although the codes are assumed herein to be orthogonal, the inventionmay be applied to nearly orthogonal (quasi-orthogonal) codes as well.

According to a first embodiment of the receiver, the vector de-spreaderis operable to despread the received spread signal using the pluralityof non data-carrying orthogonal codes to produce a plurality of nondata-carrying despread signals and to despread the received spreadsignal using the at least one data-carrying orthogonal code to produceat least one data-carrying despread signal. Further, according to thefirst embodiment, the linear combiner is operable to apply the pluralityof complex-valued combining weights to the plurality of nondata-carrying despread signals to produce a plurality of complex-valuedadjusted non data-carrying despread signals and to combine the pluralityof complex-valued adjusted non data-carrying despread signals with theat least one data-carrying despread signal to produce the data symbolswith the burst noise substantially removed. That is, in the firstembodiment, the signals are first despread, and then the despreadsignals are combined to remove the interference.

According to a second embodiment of the receiver, the vector de-spreaderis operable to despread the received spread signal using the pluralityof non data-carrying orthogonal codes to produce a plurality of nondata-carrying despread signals and to despread the received spreadsignal using the at least one data-carrying orthogonal code to produceat least one data-carrying despread signal. Further, according to thesecond embodiment, the linear combiner is operable to apply theplurality of complex-valued combining weights to the plurality of nondata-carrying despread signals and to the at least one data-carryingdespread signal to produce a plurality of complex-valued adjusteddespread signals and to combine the plurality of complex-valued adjusteddespread signals to produce the data symbols with the burst noisesubstantially removed. That is, in the second embodiment, the signalsare first despread, and then the despread signals are combined,including a weight assigned to the data-carrying code, to remove theinterference.

According to a third embodiment of the receiver, the linear combiner isoperable to apply the plurality of complex-valued combining weights tothe plurality of non data-carrying orthogonal codes to produce aplurality of complex-valued weighted non-data carrying orthogonal codesand to combine the plurality of complex-valued weighted non-datacarrying orthogonal codes with the at least one data-carrying orthogonalcode to produce an adapted rotated code. Further, with the thirdembodiment, the vector de-spreader is operable to despread the receivedspread signal using the adapted rotated code to produce the data symbolswith the burst noise substantially removed. That is, in the secondembodiment, the codes are first combined to produce an adapted rotatedcode, then the signals are despread using this combined code to removethe interference.

According to a fourth embodiment of the receiver, the linear combiner isoperable to apply the plurality of complex-valued combining weights tothe plurality of non data-carrying orthogonal codes and to the at leastone data-carrying orthogonal code to produce a plurality ofcomplex-valued weighted orthogonal codes and to combine the plurality ofcomplex-valued weighted orthogonal codes to produce an adapted rotatedcode. Further, with the fourth embodiment, the vector de-spreader isoperable to despread the received spread signal using the adaptedrotated code to produce the data symbols with the burst noisesubstantially removed. That is, in the second embodiment, the codes arefirst combined to produce an adapted rotated code, including a weightassigned to the data-carrying code, then the signals are despread usingthis combined code to remove the interference.

The weight computation functional block may employ at least one of leastmean square processing and least square processing, or any otheroptimization method, to calculate the plurality of complex-valuedcombining weights. The data symbols may be modulated according to, forexample, Binary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying(QPSK), 8 Quadrature Amplitude Modulation (QAM), 16 QAM, 32 QAM, 64 QAM,128 QAM, 256 QAM, 516 QAM, and 1024 QAM. The receiver may be, forexample, a multi-channel headend physical layer burst receiver, a singlechip wireless modem, a single chip Data Over Cable Service InterfaceSpecification (DOCSIS)/EuroDOCSIS cable modem, a base station receiver,a mobile receiver, a satellite earth station, a tower receiver, a highdefinition television set top box receiver, and a transceiver. Anotherembodiment of the present invention includes simply a receiver. Stillanother embodiment includes a method of operation. In addition, otheraspects, advantages, and novel features of the invention will becomeapparent from the following detailed description of the invention whenconsidered in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

A better understanding of the present invention can be obtained when thefollowing detailed description is considered in conjunction with thefollowing drawings.

FIG. 1 is a system diagram illustrating an embodiment of a cable modem(CM) communication system that is built according to the presentinvention.

FIG. 2 is a system diagram illustrating another embodiment of a CMcommunication system that is built according to the present invention.

FIG. 3A is a system diagram illustrating an embodiment of a cellularcommunication system that is built according to the present invention.

FIG. 3B is a system diagram illustrating another embodiment of acellular communication system that is built according to the presentinvention.

FIG. 4 is a system diagram illustrating an embodiment of a satellitecommunication system that is built according to the present invention.

FIG. 5A is a system diagram illustrating an embodiment of a microwavecommunication system that is built according to the present invention.

FIG. 5B is a system diagram illustrating an embodiment of apoint-to-point radio communication system that is built according to thepresent invention.

FIG. 6 is a system diagram illustrating an embodiment of a highdefinition (HDTV) communication system that is built according to thepresent invention.

FIG. 7 is a system diagram illustrating an embodiment of a communicationsystem that is built according to the present invention.

FIG. 8 is a system diagram illustrating another embodiment of acommunication system that is built according to the present invention.

FIG. 9 is a system diagram illustrating an embodiment of a cable modemtermination system (CMTS) system that is built according to the presentinvention.

FIG. 10 is a system diagram illustrating an embodiment of a burstreceiver system that is built according to the present invention.

FIG. 11 is a system diagram illustrating an embodiment of a single chipDOCSIS/EuroDOCSIS CM system that is built according to the presentinvention.

FIG. 12 is a system diagram illustrating another embodiment of a singlechip DOCSIS/EuroDOCSIS CM system that is built according to the presentinvention.

FIG. 13 is a system diagram illustrating an embodiment of a single chipwireless modem system that is built according to the present invention.

FIG. 14 is a system diagram illustrating another embodiment of a singlechip wireless modem system that is built according to the presentinvention.

FIG. 15 is a diagram illustrating an embodiment of a vector de-spreaderthat is built according to the present invention.

FIG. 16 is a diagram illustrating an embodiment of an interferencecanceller that is built according to the present invention.

FIG. 17 is a diagram illustrating another embodiment of an interferencecanceller that is built according to the present invention.

FIG. 18 is a diagram illustrating another embodiment of an interferencecanceller that is built according to the present invention.

FIG. 19 is a diagram illustrating an embodiment of an interferencecanceller with memory that is built according to the present invention.

FIG. 20 is a diagram illustrating an embodiment of equalization withcanceller that is arranged according to the present invention.

FIG. 21 is a diagram illustrating an embodiment of Least Means Square(LMS) training of an interference canceller according to the presentinvention.

FIG. 22A is a diagram illustrating an embodiment of signaltransformation according to the present invention.

FIG. 22B is a diagram illustrating another embodiment of signaltransformation according to the present invention.

FIG. 23 is an operational flow diagram illustrating an embodiment of aninterference cancellation method that is performed according to thepresent invention.

FIG. 24 is an operational flow diagram illustrating another embodimentof an interference cancellation method that is performed according tothe present invention.

FIG. 25 is an operational flow diagram illustrating an embodiment of anunused code selection method that is performed according to the presentinvention.

FIG. 26 is an operational flow diagram illustrating an embodiment of anS-CDMA interference cancellation method that is performed according tothe present invention.

FIG. 27 is an operational flow diagram illustrating another embodimentof an interference cancellation method that is performed according tothe present invention.

FIG. 28 is a diagram illustrating an embodiment of a spectrum ofnarrowband interference that may be addressed and overcome whenpracticing via the present invention.

FIG. 29 is a diagram illustrating an embodiment of a spectrum of adaptedcode showing null at a location of interference that may be achievedwhen practicing via the present invention.

FIG. 30A is a diagram illustrating an embodiment of a receivedconstellation before interference has been cancelled when practicing viathe present invention.

FIG. 30B is a diagram illustrating an embodiment of a receivedconstellation after interference has been cancelled when practicing viathe present invention.

FIG. 31 is a flow chart illustrating an embodiment of a method of thepresent invention for removing burst noise from a received spreadsignal.

FIG. 32 is a graph illustrating the manner in which burst noise mayaffect a portion of the chips of a symbol of the spread signal,referenced with respect to chips of an orthogonal code (128 chips persymbol in this example).

FIG. 33 is a flow chart illustrating a first embodiment of the method ofFIG. 31 that may be implemented with the structure of FIG. 16.

FIG. 34 is a flow chart illustrating a second embodiment of the methodof FIG. 31 that may be implemented with the structure of FIG. 17.

FIG. 35 is a diagram illustrating an embodiment of an interferencecanceller that is built according to the present invention illustratinga third or fourth embodiment of the method of FIG. 31.

FIG. 36 is a flow chart illustrating a third embodiment of the method ofFIG. 32 that may be implemented with the structure of FIG. 35.

FIG. 37 is a flow chart illustrating a fourth embodiment of the methodof FIG. 32 that may be implemented with the structure of FIG. 35.

FIG. 38 is a chart illustrating the property in which an adapted rotatedcode is virtually zero during the period in which the burst noise ispresent in FIG. 32, thereby allowing the adapted rotated code to beemployed to attenuate the burst noise illustrated in FIG. 32.

DETAILED DESCRIPTION OF THE INVENTION

The present invention provides a solution for interference cancellationfor communication systems (where a medium is used by one user or ifshared among many users). More specifically, the present invention isapplicable within code-division multiple access (CDMA) communicationsystems, as well as synchronous code-division multiple access (S-CDMA)communication systems. One particular type of S-CDMA communicationsystem that may benefit from the present invention is the Data OverCable Service Interface Specifications (DOCSIS) version 2.0 S-CDMA thatis operable for communication systems. The present invention presents asolution that provides for cancellation of interference in any suchcommunication systems. The interference cancellation may be viewed asbeing directed primarily towards the type of interference that occupiesa small number of signal dimensions. Examples of such types ofinterference include narrowband interference (herein also referred to asingress) or impulse/burst noise. The present invention also provides asolution where it may operate in the presence of simultaneous narrowbandinterference and impulse/burst noise and to substantially eliminate themboth.

The present invention provides an approach for interference cancellationthat provides a number of benefits including a completely linear methodof canceling ingress that employs no DFE (Decision Feedback Equalizer)or SIC (Successive Ingress Cancellation). This approach can cancelwideband ingress and may be implemented in a relatively simple andefficient structure. Moreover, the present invention may be combinedrelatively easily, given its simple and efficient structure, with othermethods and systems that may assist in the interference cancellation.

The present invention uses a linear combination of the unused dimensions(for example, the unused orthogonal codes) to cancel the interference.This may be done after the de-spreader, or, equivalently, as part of thede-spreading process. There is no appreciable inter-code interferenceintroduced and no decision errors are made as part of this process. Theallocation of unused codes reduces the capacity of the system by a smallamount; for example, to 120/128 of the original capacity for a system of128 codes where 8 unused codes are employed. The term “codes” isgenerally used herein to describe orthogonal codes. In some embodiments,these codes are orthonormal as well. Quasi-orthogonal codes may also beused. It is noted here that the specific examples of 120 active codesand 8 unused codes in a system having 128 available codes is exemplary.Clearly, other embodiments may be employed (having different numbers ofcodes—both different numbers of used and unused) without departing fromthe scope and spirit of the invention.

From certain perspectives, the present invention may be viewed as beingan extension of sub-dimensional modulation that is employed for TDMA(Time Division Multiple Access), and it may then be extended to S-CDMA(Synchronous Code Division Multiple Access). To begin this discussion,we consider an example case where 120 active codes are used andnarrowband ingress (narrowband interference) is present. (The sameprinciples will apply to burst noise cancellation as to narrowbandingress cancellation.) The remaining 8 codes are then transmitted astrue zero symbols. At the receiver, the unused 8 codes are received withsamples of the narrowband ingress. This information may then be used tocancel the ingress in the same manner that is used in sub-dimensionalmodulation. The 8 unused codes are considered as extra dimensions andthe receiver is trained based on this information.

We continue on with 128 code example (8 of the codes being unused). Byemploying the 8 unused spreading codes, linear combinations of these 8codes may be added to any other code that is transmitted from atransmitter to a receiver, and the resulting composite waveform can beused for de-spreading that one transmitted code. Since, ideally, nosignal is present on any of these 8 codes (being zero codes), little orno inter-code interference from the transmitters will be induced. InAWGN (Additive White Gaussian Noise), the inclusion of these additionalde-spreading codes serves to merely increase the noise at the decisionslicer for the code of interest. However, in the presence ofinterference, such as narrowband interference, some combination of thesecodes in the de-spreading process is operable to reduce the ingressenough to overcome the increased AWGN and make it worthwhile.Heuristically, the unused codes may correlate somewhat with theinterference, and if so, may be used to “subtract” some component of theinterference at the decision slicer, as a type of noise canceller.

The similarity of this approach is somewhat analogous to thefunctionality of an ICF (Ingress Cancellation Filter). In certainembodiments, a TDMA-only, CDMA-only, and/or a TDMA/CDMA burst receivermay be leveraged for this S-CDMA application, including the use of theTrench method or derivative that is mentioned below.

It is noted that the approach to reducing the interference power at thedecision point using only the unused codes may not be optimal in certainembodiments, since using codes with data carried on them may offer someinterference rejection overcoming the introduction of the transmitterinter-code interference. It is also noted that including “in use” codeswould actually provide benefit in practical situations.

It is also noted that using a strict LMS (Least Mean Square) type ofapproach to converging 128 taps to the desired de-spreading code mayprove laborious (in terms of requiring many iterations). This approachdoes not focus on just the unused codes, and has far more degrees offreedom than just finding 8 coefficients for the weightings of the 8unused codes. If LMS is used, constraints may be employed to reduce thenumber of degrees of freedom.

Analysis has shown that indeed there is great similarity in theformulation and solution of the optimal taps in the monic filter of theICF for ingress cancellation, and in this application of unusedspreading codes in ingress cancellation for S-CDMA. Both can beformulated in a LS (Least Squares) type of problem, with the ensuingtypical solution taking form.

One advantage that may arise for the TDMA ICF case is when the matrix tobe inverted is Toeplitz (a Toeplitz matrix in addition a matrix in whichall the elements are the same along any diagonal that slopes fromnorthwest to southeast), and thus admits significant computationaladvantages, such as discovered by William Trench.

However, in the S-CDMA formulation with unused codes, the matrix to beinverted is not Toeplitz, and the Trench approach does not apply. Therestill may be some computational advantages due to the underlyingconstruction of the matrix to be inverted, but it may be that only thetraditional numerical methods such as Cholesky decomposition mayintroduce simplification. The matrix to be inverted is at least of theform C*RC, where C=128×8, with each column orthogonal with the others (asub-matrix of a Unitary matrix), and R=128×128 and is Toeplitz, and *stands for complex conjugate.

One implementation of the present invention may be described as shownbelow.

Let R=R_(m,n)=E {r*(m) r(n)}, where r(n) are noise and ingress samples,containing little or no signal. R is 128×128, where the samplescorrespond to the noise in the chips of a spreading interval. Thesymbol * denotes complex conjugate. This is the noise (or noise plusinterference) covariance matrix.

Let C=[c1 c2 c3 c4 c5 c6 c7 c8], where ci=column of 128 chips of i^(th)unused spreading code. C is 128 rows by 8 columns.

Let:

dsc_(opt)=optimal de-spreading code for code-of-interest c_(k), (writtenalternatively below)dsc _(opt) =c _(k) +w ₁ c1+w ₂ c2+ . . . +w ₈ c8,

where dsc_(opt) is a column vector with 128 components, and w_(i) arescalar weighting coefficients.

Thus, dsc_(opt)=[1 w₁ w₂ . . . w₈][c_(k) c1 c2 . . . c8], and we can seethat the optimal solution for the S-CDMA case indeed has a form similarto the monic filter in the TDMA ICF solution.

Let w_(opt)=[w₁ w₂ . . . w₈] which provides the optimal de-spreading forsignaling with the k^(th) spreading code.

It can be shown that w_(opt)=−[C^(T)RC]⁻¹[C^(T)R]c_(k), where all thevectors, matrices, and notation are as defined above.

It is noted that with the ICF in TDMA,

w_(opt) = −[R]⁻¹ [r 1   r 2   r 3   …   r 16]^(T), where$R = \begin{bmatrix}{r\; 0} & {r\; 1} & {r\; 2} & \ldots & {r\; 16} \\{r\; 1^{*}} & {r\; 0} & {r\; 1} & \ldots & {r\; 15} \\\vdots & \; & \; & \ddots & \vdots \\{r\; 16^{*}} & {r\; 15^{*}} & \ldots & {r\; 1^{*}} & {r\; 0}\end{bmatrix}$

Another characteristic of the present invention is that the complexityof the method with the LS approach is high for large matrices comparedto the ICF and Trench method with TDMA. However, this complexity ismitigated by the reduced matrix size resulting from the sub-spaceprojection approach described here and below. From certain perspectives,the S-CDMA approach with unused codes may be viewed as not lendingitself to the computational efficiencies of Trench approach or itsderivatives.

However, using an LMS tap update approach has been found to work. Theweights of the unused spreading codes, wi, are updated in LMS fashion.In this method, the de-spreader for the code of interest is input to adecision slicer. The resulting complex error is computed. Similarly,each unused spreading code has its corresponding de-spreading operating.The result of the de-spreader corresponding to an unused spreading codeis “signal,” just as the “signal” rests in the various shift registersin the conventional FIR (Finite Impulse Response) filter in the normalLMS. The de-spread outputs for each of the unused codes is multiplied bythe error vector, and then multiplied by a step size factor “−mu” andadded to the existing tap weight, to update the tap weight.

With 8 unused spreading codes, there is a set of 8 tap weights for eachused spreading code. Thus, with the 8 unused spreading codes, there are8×120 tap weights to iterate.

This approach is much less computationally intensive than using anLS-based. The S-CDMA LMS approach introduces little additionalcomputation when compared to operating the de-spreading codesthemselves. This approach is beneficial for a number of reasons. It iscomputationally acceptable, in that, it limits the number of taps toonly 8 (in our continuing example of 8 unused spreading codes), whichalmost certainly provides much more rapid convergence and less “tapnoise.” In addition, it eliminates from the search for optimalcoefficients in the de-spreader those codes that are known to correspondto used codes.

FIG. 1 is a system diagram illustrating an embodiment of a CMcommunication system 100 that is built according to the presentinvention. The CM communication system includes a number of CMs (shownas a CM user A#1 111, a CM user #2 115, . . . , and a CM user #n 121)and a CMTS 130. The CMTS 130 is a component that exchanges digitalsignals with CMs on a cable network.

Each of a number of CM users, shown as the CM user #1 111, the CM user#2 115, . . . , and the CM user #n 121, is able to communicativelycouple to a CM network segment 199. A number of elements may be includedwithin the CM network segment 199. For example, routers, splitters,couplers, relays, and amplifiers may be contained within the CM networksegment 199 without departing from the scope and spirit of theinvention.

The CM network segment 199 allows communicative coupling between a CMuser and a cable headend transmitter 120 and/or a CMTS 130. In someembodiments, a cable CMTS is in fact contained within a headendtransmitter. In other embodiments, the functionality of the cable CMTSand the headend transmitter are represented as two distinct functionalblocks so that their respective contribution may be more easilyappreciated and understood. This viewpoint is shown in the situationwhere the CMTS 130 is pictorially shown as being located externally to acable headend transmitter 120. In the more common representation andimplementation, a CMTS 135 is located within the cable headendtransmitter 120. The combination of a CMTS and a cable headendtransmitter may be referred to as being the “cable headend transmitter”which supports the CMTS functionality. The CMTS 130 may be located at alocal office of a cable television company or at another location withina CM communication system. In the following description, the CMTS 130 isused for illustration; yet, the same functionality and capability asdescribed for the CMTS 130 may equally apply to embodiments thatalternatively employ the CMTS 135. The cable headend transmitter 120 isable to provide a number of services including those of audio, video,telephony, local access channels, as well as any other service known inthe art of cable systems. Each of these services may be provided to theone or more CM users 111, 115, . . . , and 121.

In addition, through the CMTS 130, the CM users 111, 115, . . . , 121are able to transmit and receive data from the Internet, . . . , and/orany other network to which the CMTS 130 is communicatively coupled. Theoperation of a CMTS, at the cable-provider's head-end, may be viewed asproviding many of the same functions provided by a digital subscriberline access multiplexer (DSLAM) within a digital subscriber line (DSL)system. The CMTS 130 takes the traffic coming in from a group ofcustomers on a single channel and routes it to an Internet ServiceProvider (ISP) for connection to the Internet, as shown via the Internetaccess. At the head-end, the cable providers will have, or lease spacefor a third-party ISP to have, servers for accounting and logging,dynamic host configuration protocol (DHCP) for assigning andadministering the Internet protocol (IP) addresses of all the cablesystem's users, and typically control servers for a protocol called DataOver Cable Service Interface Specifications (DOCSIS), the major standardused by U.S. cable systems in providing Internet access to users.

The downstream information flows to all of the connected CM users 111,115, . . . , 121; this may be viewed to be in a manner that is similarto that manner within an Ethernet network. The individual networkconnection, within the CM network segment 199, decides whether aparticular block of data is intended for it or not. On the upstreamside, information is sent from the CM users 111, 115, . . . , 121 to theCMTS 130; on this upstream transmission, the users within the CM users111, 115, . . . , 121 to whom the data is not intended do not see thatdata at all. As an example of the capabilities provided by a CMTS, theCMTS will enable as many as 1,000 users to connect to the Internetthrough a single 6 MHz channel. Since a single channel is capable of30-40 megabits per second of total throughput, this means that users maysee far better performance than is available with standard dial-upmodems. Embodiments implementing the present invention are describedbelow and in the various Figures that show the data handling and controlwithin one or both of a CM and a CMTS within a CM system that operatesby employing S-CDMA (Synchronous Code Division Multiple Access).

The CMs of the CM users 111, 115, . . . , 121 and the CMTS 130communicate synchronization information to one another to ensure properalignment of transmission from the CM users 111, 115, . . . , 121 to theCMTS 130. This is where the synchronization of the S-CDMA communicationsystems is extremely important. When a number of the CMs all transmittheir signals at a same time such that these signals are received at theCMTS 130 on the same frequency and at the same time, they must all beable to be properly de-spread and decoded for proper signal processing.

Each of the CMs users 111, 115, . . . , 121 is located a respectivetransmit distance from the CMTS 130. In order to achieve optimumspreading diversity and orthogonality for the CMs users 111, 115, . . ., 121 to transmission of the CMTS 130, each of the CM transmissions mustbe synchronized so that it arrives, from the perspective of the CMTS130, synchronous with other CM transmissions. In order to achieve thisgoal, for a particular transmission cycle, each of the CMs 111, 115, . .. , 121 will typically transmit to the CMTS 130 at a respectivetransmission time, which will likely differ from the transmission timesof other CMs. These differing transmission times will be based upon therelative transmission distance between the CM and the CMTS 130. Theseoperations may be supported by the determination of the round tripdelays (RTPs) between the CMTS 130 and each supported CM. With theseRTPs determined, the CMs may then determine at what point to transmittheir S-CDMA data so that all CM transmissions will arrive synchronouslyat the CMTS 130.

The present invention enables interference cancellation within the CMTS130, as shown in a functional block 131. The present invention may alsobe implemented to support interference cancellation within any one ofthe CMs 111, 115, . . . , 121; the particular implementation ofinterference cancellation is shown as a functional block 122 within theCM 122, yet it is understood that the interference cancellationfunctionality may also be supported within the other CMs as well. TheCMTS 130 receives an input spread signal and is operable to performdispreading and interference cancellation according to the presentinvention. The CMTS 130 is operable to employ a linear combiner thatuses as inputs complex valued combining weights to the particular codesthat are selectively used to assist in the interference cancellation ofone of the de-spread signals that is de-spread from the input spreadsignal that the CMTS 130 receives. Alternatively, the present inventionmay be viewed as employing at least one of an adapted code and anadapted code matrix to perform interference cancellation according tothe present invention.

FIG. 2 is a system diagram illustrating another embodiment of a CMcommunication system 200 that is built according to the presentinvention. From certain perspectives, the FIG. 2 may be viewed as acommunication system allowing bi-directional communication betweencustomer premise equipment (CPE) 240 and a network. In some embodiments,the CPE 240 is a personal computer or some other device allowing a userto access an external network. The network may be a wide area network(WAN) 280, or alternatively, the Internet 290 itself. For example, theCM communication system 200 is operable to allow Internet protocol (IP)traffic to achieve transparent bidirectional transfer between aCMTS-network side interface (CMTS-NSI: viewed as being between the CMTS230 and the Internet 290) and a CM to CPE interface (CMCI: viewed asbeing between the CM 210 and the CPE 240).

The WAN 280, and/or the Internet 290, is/are communicatively coupled tothe CMTS 230 via a CMTS-NSI. The CMTS 230 is operable to support theexternal network termination, for one or both of the WAN 280 and theInternet 290. The CMTS 230 includes a modulator and a demodulator tosupport transmitter and receiver functionality to and from a CM networksegment 299. The receiver functionality within the CMTS 230 is operableto support interference cancellation functionality 231 according to thepresent invention. It is also noted that there may be embodiment wherethe CM 210 is also operable to support interference cancellationfunctionality using the present invention, as shown by a functionalblock 211. Implementing interference cancellation in the transmitterprevents noise enhancement that occurs when interference cancellation isperformed in the receiver.

A number of elements may be included within the CM network segment 299.For example, routers, splitters, couplers, relays, and amplifiers may becontained within the CM network segment 299 without departing from thescope and spirit of the invention. The CM network segment 299 allowscommunicative coupling between a CM user and the CMTS 230. The FIG. 2shows just one of many embodiments where the interference cancellation,performed according to the present invention, may be performed toprovide for improved operation within a communication system.

FIG. 3A is a system diagram illustrating an embodiment of a cellularcommunication system 300A that is built according to the presentinvention. A mobile transmitter 310 has a local antenna 311. The mobiletransmitter 310 may be any number of types of transmitters including acellular telephone, a wireless pager unit, a mobile computer havingtransmit functionality, or any other type of mobile transmitter. Themobile transmitter 310 transmits a signal, using its local antenna 311,to a base station receiver 340 via a wireless communication channel. Thebase station receiver 340 is communicatively coupled to a receivingwireless tower 349 to be able to receive transmission from the localantenna 311 of the mobile transmitter 310 that have been communicatedvia the wireless communication channel. The receiving wireless tower 349communicatively couples the received signal to the base station receiver340.

The base station receiver 340 is then able to support interferencecancellation functionality according to the present invention, as shownin a functional block 341, on the received signal. The FIG. 3A shows yetanother of the many embodiments where the interference cancellation,performed according to the present invention, may be performed toprovide for improved operation within a communication system.

FIG. 3B is a system diagram illustrating another embodiment of acellular communication system that is built according to the presentinvention. From certain perspectives, the FIG. 3B may be viewed as beingthe reverse transmission operation of the cellular communication system300B of the FIG. 3A. A base station transmitter 320 is communicativelycoupled to a transmitting wireless tower 321. The base stationtransmitter 320, using its transmitting wireless tower 321, transmits asignal to a local antenna 339 via a wireless communication channel. Thelocal antenna 339 is communicatively coupled to a mobile receiver 330 sothat the mobile receiver 330 is able to receive transmission from thetransmitting wireless tower 321 of the base station transmitter 320 thathave been communicated via the wireless communication channel. The localantenna 339 communicatively couples the received signal to the mobilereceiver 330. It is noted that the mobile receiver 330 may be any numberof types of transmitters including a cellular telephone, a wirelesspager unit, a mobile computer having transmit functionality, or anyother type of mobile transmitter.

The mobile receiver 330 is then able to support interferencecancellation functionality according to the present invention, as shownin a functional block 331, on the received signal. The FIG. 3B shows yetanother of the many embodiments where the interference cancellationfunctionality, performed according to the present invention, may beperformed to provide for improved operation within a communicationsystem.

It is also noted that the embodiments described above within the FIGS.3A and 3B may operate in conjunction within a single communicationsystem. That is to say, a mobile unit (that supports both transmit andreceive functionality) may be implemented to support interferencecancellation functionality during receipt of signals while the basestation device (that supports both transmit and receive functionality)may also be implemented to support interference cancellationfunctionality during receipt of signals. This way, both devices areoperable to support the interference cancellation functionalityaccording to the present invention at both ends of the communicationlink. This dual-end interference cancellation functionality is also truewithin other of the various embodiments described herein that illustrateboth ends of a communication link.

It is further noted that the embodiments described above within theFIGS. 3A and 3B may operate in conjunction within a single communicationsystem from yet another perspective. A mobile transmitter may beimplemented to support interference cancellation functionality duringsignal processing and transmission of its signals to a base stationreceiver. Similarly, a base station transmitter may be implemented tosupport interference cancellation functionality during signal processingand transmission of its signals to a mobile unit receiver. This may beperformed, at least in part, by adjusting a transmitted spectrum to meeta desired spectral mask. It may be desirable to attenuate certainportions of the spectrum using the subspace canceller. In theseapplications the canceller is predominantly located in the transmitter.Further detail of this interference cancellation within a transmitterdevice is presented below. This adjusting of a transmitted spectrum tomeet a desired spectral mask may be performed within any of the variousembodiments that include a transmitter that transmits a signal to areceiver according to the present invention.

FIG. 4 is a system diagram illustrating an embodiment of a satellitecommunication system 400 that is built according to the presentinvention. A transmitter 420 is communicatively coupled to a wirednetwork 410. The wired network 410 may include any number of networksincluding the Internet, proprietary networks, and other wired networks.The transmitter 420 includes a satellite earth station 451 that is ableto communicate to a satellite 453 via a wireless communication channel.The satellite 453 is able to communicate with a receiver 430. Thereceiver 430 is also located on the earth. Here, the communication toand from the satellite 453 may cooperatively be viewed as being awireless communication channel, or each of the communication to and fromthe satellite 453 may be viewed as being two distinct wirelesscommunication channels.

For example, the wireless communication “channel” may be viewed as notincluding multiple wireless hops in one embodiment. In otherembodiments, the satellite 453 receives a signal received from thesatellite earth station 451, amplifies it, and relays it to the receiver430; the receiver 430 may include terrestrial receivers such assatellite receivers, satellite based telephones, and satellite basedInternet receivers, among other receiver types. In the case where thesatellite 453 receives a signal received from the satellite earthstation 451, amplifies it, and relays it, the satellite 453 may beviewed as being a “transponder.” In addition, other satellites may existthat perform both receiver and transmitter operations. In this case,each leg of an up-down transmission via the wireless communicationchannel would be considered separately. The wireless communicationchannel between the satellite 453 and a fixed earth station would likelybe less time-varying than the wireless communication channel between thesatellite 453 and a mobile station.

In whichever embodiment, the satellite 453 communicates with thereceiver 430. The receiver 430 may be viewed as being a mobile unit incertain embodiments (employing a local antenna 412); alternatively, thereceiver 430 may be viewed as being a satellite earth station 452 thatmay be communicatively coupled to a wired network in a similar mannerthat the satellite earth station 451, within the transmitter 420,communicatively couples to a wired network. In both situations, thereceiver 430 is able to support interference cancellation functionality,as shown in a functional block 431, according to the present invention.For example, the receiver 430 is able to perform interferencecancellation, as shown in a functional block 431, on the signal receivedfrom the satellite 453. The FIG. 4 shows yet another of the manyembodiments where the interference cancellation, performed according tothe present invention, may be performed to provide for improved receiverperformance.

FIG. 5A is a system diagram illustrating an embodiment of a microwavecommunication system 500A that is built according to the presentinvention. A tower transmitter 511 includes a wireless tower 515. Thetower transmitter 511, using its wireless tower 515, transmits a signalto a tower receiver 512 via a wireless communication channel. The towerreceiver 512 includes a wireless tower 516. The wireless tower 516 isable to receive transmissions from the wireless tower 515 that have beencommunicated via the wireless communication channel. The tower receiver512 is then able to support interference cancellation functionality, asshown in a functional block 533. The FIG. 5A shows yet another of themany embodiments where interference cancellation, performed according tothe present invention, may be performed to provide for improved receiverperformance.

FIG. 5B is a system diagram illustrating an embodiment of apoint-to-point radio communication system 500B that is built accordingto the present invention. A mobile unit 551 includes a local antenna555. The mobile unit 551, using its local antenna 555, transmits asignal to a local antenna 556 via a wireless communication channel. Thelocal antenna 556 is included within a mobile unit 552. The mobile unit552 is able to receive transmissions from the mobile unit 551 that havebeen communicated via the wireless communication channel. The mobileunit 552 is then able to support interference cancellationfunctionality, as shown in a functional block 553, on the receivedsignal. The FIG. 5B shows just yet another of the many embodiments whereinterference cancellation, performed according to the present invention,may be performed to provide for improved receiver performance.

FIG. 6 is a system diagram illustrating an embodiment of a highdefinition (HDTV) communication system 600 that is built according tothe present invention. An HDTV transmitter 610 includes a wireless tower611. The HDTV transmitter 610, using its wireless tower 611, transmits asignal to an HDTV set top box receiver 620 via a wireless communicationchannel. The HDTV set top box receiver 620 includes the functionality toreceive the wireless transmitted signal. The HDTV set top box receiver620 is also communicatively coupled to an HDTV display 630 that is ableto display the demodulated and decoded wireless transmitted signalsreceived by the HDTV set top box receiver 620.

The HDTV set top box receiver 620 is then able to support interferencecancellation functionality, as shown in a functional block 623 toprovide for improved receiver performance. The FIG. 6 shows yet anotherof the many embodiments where interference cancellation, performedaccording to the present invention, may be performed to provide forimproved receiver performance.

FIG. 7 is a system diagram illustrating an embodiment of a communicationsystem that is built according to the present invention. The FIG. 7shows communicative coupling, via a communication channel 799, betweentwo transceivers, namely, a transceiver 701 and a transceiver 702. Thecommunication channel 799 may be a wire line communication channel or awireless communication channel.

Each of the transceivers 701 and 702 includes a transmitter and areceiver. For example, the transceiver 701 includes a transmitter 749and a receiver 740; the transceiver 702 includes a transmitter 759 and areceiver 730. The receivers 740 and 730, within the transceivers 701 and702, respectively, are each operable to support interferencecancellation functionality according to the present invention. This willallow improved signal processing for both of the transceivers 701 and702. For example, the receiver 740, within the transceiver 701, is ableto support interference cancellation functionality, as shown in afunctional block 741, on a signal received from the transmitter 759 ofthe transceiver 702. Similarly, the receiver 730, within the transceiver702, is able to support interference cancellation functionality, asshown in a functional block 731, on a signal received from thetransmitter 749 of the transceiver 701.

If desired in certain embodiments, the transmitters 749 and 759, withinthe transceivers 701 and 702, respectively, are each operable to supportinterference cancellation functionality according to the presentinvention. This will also allow improved signal processing for both ofthe transceivers 701 and 702. For example, the transmitter 749, withinthe transceiver 701, is able to support interference cancellationfunctionality, as shown in a functional block 748, on a signal that isto be transmitted from the transmitter 759 of the transceiver 702.Similarly, the transmitter 759, within the transceiver 702, is able tosupport interference cancellation functionality, as shown in afunctional block 758, on a signal that is to be transmitted from thetransmitter 759 of the transceiver 702.

This interference cancellation functionality, within the transmitters749 and 759, respectively, may be performed, at least in part, byadjusting a transmitted spectrum to meet a desired spectral maskaccording to the present invention. The FIG. 7 shows yet another of themany embodiments where interference cancellation, performed according tothe present invention, may be performed to provide for improvedperformance.

FIG. 8 is a system diagram illustrating another embodiment of acommunication system 800 that is built according to the presentinvention. The FIG. 8 shows communicative coupling, via a communicationchannel 899, between a transmitter 849 and a receiver 830. Thecommunication channel 899 may be a wire line communication channel or awireless communication channel. The receiver 830 is operable to supportinterference cancellation, as shown in a functional block 831, accordingto the present invention. The FIG. 8 shows yet another of the manyembodiments where interference cancellation, performed according to thepresent invention, may be performed to provide for improved performance.

In certain embodiments, the transmitter 849 is also operable to supportinterference cancellation, as shown in a functional block 848, accordingto the present invention. This interference cancellation functionality,within the transmitter 849, may be performed, at least in part, byadjusting a transmitted spectrum to meet a desired spectral maskaccording to the present invention. For example, the interferencecancellation functionality of the present invention may be located atthe transmitter 849, the receiver 830, or partly in each (as shown bythe functional blocks 848 and 831). In many of the various embodimentsdescribed herein, the interference cancellation functionality has beenlocated in a receiver of a communication system. The following describesone embodiment of how the interference cancellation functionality may belocated at the transmitter. The unused codes, instead of being modulatedat the transmitter with zero symbols, are modulated with a linearcombination of the desired signals from the used codes. In the case ofnarrowband interference, the resulting transmitted signal will have anull on the interferer.

FIG. 9 is a system diagram illustrating an embodiment of a CMTS system900 that is built according to the present invention. The CMTS system900 includes a CMTS medium access controller (MAC) 930 that operateswith a number of other devices to perform communication from one or moreCMs to a WAN 980. The CMTS MAC 930 may be viewed as providing thehardware support for MAC-layer per-packet functions includingfragmentation, concatenation, and payload header suppression that allare able to offload the processing required by a system centralprocessing unit (CPU) 972. This will provide for higher overall systemperformance. In addition, the CMTS MAC 930 is able to provide supportfor carrier class redundancy via timestamp synchronization across anumber of receivers, shown as a receiver 911, a receiver 911, and areceiver 913 that are each operable to receive upstream analog inputs.In certain embodiments, each of the receivers 911, 912, and 913 are dualuniversal advanced TDMA/CDMA (Time Division Multiple Access/CodeDivision Multiple Access) PHY-layer burst receivers. That is top say,each of the receivers 911, 912, and 913 includes at least one TDMAreceive channel and at least one CDMA receive channel; in this case,each of the receivers 911, 912, and 913 may be viewed as beingmulti-channel receivers.

In addition, the CMTS MAC 930 may be operated remotely with arouting/classification engine 979 that is located externally to the CMTSMAC 930 for distributed CMTS applications including mini fiber nodeapplications. Moreover, a Standard Programming Interface (SPI) masterport may be employed to control the interface to the receivers 911, 912,and 913 as well as to a downstream modulator 920.

The CMTS MAC 930 may be viewed as being a highly integrated CMTS MACintegrated circuit (IC) for use within the various DOCSIS and advancedTDMA/CDMA physical layer (PHY-layer) CMTS products. The CMTS MAC 930 mayemploy hardware engines for upstream and downstream paths. The upstreamprocessor design is segmented and uses two banks of Synchronous DynamicRandom Access Memory (SDRAM) to minimize latency on internal buses. Thetwo banks of SDRAM used by the upstream processor are shown as upstreamSDRAM 975 (operable to support keys and reassembly) and SDRAM 976(operable to support Packaging, Handling, and Storage (PHS) and outputqueues). The upstream processor performs Data Encryption Standard (DES)decryption, fragment reassembly, de-concatenation, payload packetexpansion, packet acceleration, upstream Management Information Base(MIB) statistic gathering, and priority queuing for the resultantpackets. Each output queue can be independently configured to outputpackets to either a Personal Computer Interface (PCI) or a Gigabit MediaIndependent Interface (GMII). DOCSIS MAC management messages andbandwidth requests are extracted and queued separately from data packetsso that they are readily available to the system controller.

The downstream processor accepts packets from priority queues andperforms payload header suppression, DOCSIS header creation, DESencryption, Cyclic Redundancy Check (CRC) and Header Check Sequence (ofthe DOCSIS specification), Moving Pictures Experts Group (MPEG)encapsulation and multiplexing, and timestamp generation on the in-banddata. The CMTS MAC 930 includes an out-of-band generator and TDMAPHY-layer (and/or CDMA PHY-layer) interface that supports communicationwith a CM device's out-of-band receiver for control of power managementfunctions. The downstream processor will also use SDRAM 977 (operable tosupport PHS and output queues). The CMTS MAC 930 may be configured andmanaged externally via a PCI interface and a PCI bus 971.

Each of the receivers 911, 912, and 913 is operable to supportinterference cancellation functionality. For example, the receiver 911is operable to support interference cancellation functionality, as shownin a functional block 991; the receiver 912 is operable to supportinterference cancellation functionality, as shown in a functional block992; and the receiver 913 is operable to support interferencecancellation functionality, as shown in a functional block 993. The FIG.9 shows yet another embodiment in which interference cancellation may beperformed according to the present invention. Any of the functionalityand operations described in the other embodiments may be performedwithin the context of the CMTS system 900 without departing from thescope and spirit of the invention.

FIG. 10 is a system diagram illustrating an embodiment of a burstreceiver system 1000 that is built according to the present invention.The burst receiver system 1000 includes at least one multi-channelreceiver 1010. The multi-channel receiver 1010 is operable to receive anumber of upstream analog inputs that are transmitted from CMs. Theupstream analog inputs may be in the form of either TDMA (Time DivisionMultiple Access) and/or CDMA (Code Division Multiple Access) format. Anumber of receive channels may be included within the multi-channelreceiver 1010.

For example, the multi-channel receiver 1010 is operable to support anumber of TDMA receive channels 1020 (shown as TDMA signal 1 and TDMAsignal 2) and to support interference cancellation functionality, asshown in a functional block 1021, for those received TDMA signals. Themulti-channel receiver 1010 is operable to support a number of TDMAreceive channels 1030 (shown as CDMA signal 1 and CDMA signal 2) and tosupport interference cancellation functionality, as shown in afunctional block 1031, for those received CDMA signals. Genericallyspeaking, the multi-channel receiver 1010 is operable to support anumber of receive channels 1040 (shown as received signal 1 and receivedsignal 2) and to support interference cancellation functionality, asshown in a functional block 1041, for those received signals. Themulti-channel receiver 1010 of the FIG. 10 is operable to interface witha CMTS MAC. The burst receiver system 1000 may include a number ofmulti-channel receivers that are each operable to interface with theCMTS MAC.

In certain embodiments, the multi-channel receiver 1010 provides anumber of various functionalities. The multi-channel receiver 1010 maybe a universal headend advanced TDMA PHY-layer QPSK/QAM (QuadraturePhase Shift Keying/Quadrature Amplitude Modulation) burst receiver; themulti-channel receiver 1010 also include functionality to be a universalheadend advanced CDMA PHY-layer QPSK/QAM burst receiver; or themulti-channel receiver 1010 also include functionality to be a universalheadend advanced TDMA/CDMA PHY-layer QPSK/QAM burst receiver offeringboth TDMA/CDMA functionality. The multi-channel receiver 1010 may beDOCSIS/EuroDOCSIS based, IEEE 802.14 compliant. The multi-channelreceiver 1010 may be adaptable to numerous programmable demodulationincluding BPSK (Binary Phase Shift Keying), QPSK, and/or8/16/32/64/128/256/516/1024 QAM. The multi-channel receiver 1010 isadaptable to support variable symbols rates as well. Other functionalitymay likewise be included to the multi-channel receiver 1010 withoutdeparting from the scope and spirit of the invention. Such variationsand modifications may be made to the communication receiver.

FIG. 11 is a system diagram illustrating an embodiment of a single chipDOCSIS/EuroDOCSIS CM system 1100 that is built according to the presentinvention. The single chip DOCSIS/EuroDOCSIS CM system 1100 includes asingle chip DOCSIS/EuroDOCSIS CM 1110 that is implemented in a very highlevel of integration and offering a very high level of performance. Acoaxial cable inputs to a DiPlexer to provide CM access to the singlechip DOCSIS/EuroDOCSIS CM system 600. The DiPlexer communicativelycouples to a CMOS (Complementary Metal Oxide Semiconductor) tuner. TheCMOS tuner may be implemented with a companion part that includes a lownoise amplifier (LNA) and performs radio frequency (RF) automatic gaincontrol (AGC). This two part solution is operable to support 64 and 256QAM and other modulations as well, e.g., BPSK, QPSK, and other QAMmodulations. These two parts operate cooperatively with the single chipDOCSIS/EuroDOCSIS CM 1110. The CMOS tuner may be operable to support anintermediate frequency (IF) output frequency range of 36-44 MHz, andspecifically support the 36.125 and 43.75 MHz center frequencies for thePhase Alteration Line (PAL) and National Television System Committee(NTSC) standards. Also, the CMOS tuner and the LNA and RF AGC are DOCSISand EuroDOCSIS standard supportable.

However, it is also noted that the CMOS tuner is operable to performdirect RF to baseband (BB) frequency transformation without requiringthe IF transformation. An external bandpass Surface Acoustic Wave (SAW)filter removes the channels distant from the desired signal.

The output from the SAW filter is then passed to the single chipDOCSIS/EuroDOCSIS CM 1110. The single chip DOCSIS/EuroDOCSIS CM 1110 issupported by Synchronous Dynamic Random Access Memory (SDRAM) and Flash.In addition, the single chip DOCSIS/EuroDOCSIS CM 1110 supports bothEthernet and USB interfacing to any other devices that may exist withinthe single chip DOCSIS/EuroDOCSIS CM system 1100. The FIG. 11 shows yetanother embodiment in which interference cancellation may be performedaccording to the present invention. The interference cancellationfunctionality may be supported directly within the single chipDOCSIS/EuroDOCSIS CM 1110. The single chip DOCSIS/EuroDOCSIS CM system1100 shows an application context of yet another implementation of adevice that may perform the present invention.

FIG. 12 is a system diagram illustrating another embodiment of a singlechip DOCSIS/EuroDOCSIS CM system 1200 that is built according to thepresent invention. The single chip DOCSIS/EuroDOCSIS CM system 1200includes a single chip DOCSIS/EuroDOCSIS CM 1210 that combines an RFreceiver with an advanced QAM demodulator, an advanced QAM and S-CDMAmodulator/transmitter, a complete DOCSIS 2.0 Media Access Controller(MAC), a 200 MHz MIPS32 Communication Processor, a 16 bit, 100 MHz SDRAMinterface, 10/100 Ethernet MAC with integrated transceiver and MediaIndependent Interface (MII), and a USB 1.1 controller with integratedtransceiver.

The QAM receiver directly samples a tuner output (such as the CMOS tunerof the FIG. 6) with an 11 bit analog to digital converter (ADC) andinput AGC amplifier. The receiver digitally re-samples and demodulatesthe signal with recovered clock and carrier timing, filters andequalizes the data, and passes soft decisions to an ITU-T J.83 AnnexA/B/C compatible decoder. The receiver supports variable symbol rate4/16/32/64/128/256/1024 QAM Forward Error Correction (FEC) decoding. Thefinal received data stream is delivered in a serial MPEG-2 transportformat. All gain, clock, and carrier, acquisition and tracking loops areintegrated in the QAM receiver.

The upstream transmitter takes burst or continuous data, provides FECencoding and pre-equalization for DOCSIS applications, filters andapplies 2/4/8/16/64/256 QAM or S-CDMA modulation to the data stream,amplifies the signal through the integrated upstream power amplifier andprovides a direct 0-65 MHz analog output.

The MAC of the single chip DOCSIS/EuroDOCSIS CM 1210 includes allfeatures required for full DOCSIS 1.0, 1.1, and 2.0 compliance,including full support for baseline privacy (BPI+) encryption anddecryption. Single-user support includes four SIDS (StandardInteroperable Datalink System) in downstream, four DA perfect matchfilters, a 256 entry CAM for multicast/unicast hash filter and fourindependent upstream queues for simultaneous support of Quality ofService (QoS) and BE traffic. To enhance operational support, the MAC ofthe MAC of the single chip DOCSIS/EuroDOCSIS CM 1210 provides extendedNetwork Management MIB/Diagnostic features, as well as immediate UCC (onthe fly) using independent resets for downstream and upstream queues andboth individual queue reset/flush for upstream queues. The MAC of thesingle chip DOCSIS/EuroDOCSIS CM 1210 uses advance PROPANE™ techniquesto provide packet acceleration to significantly improve upstream channelutilization.

With the incorporation of an upstream power amplifier, the MAC of thesingle chip DOCSIS/EuroDOCSIS CM 1210 allows a complete CM to beassembled with a minimal set of external components. When used with aCMOS tuner, such as the CMOS tuner of the FIG. 11, a very low costsolution for a high performance, single user DOCSIS 2.0 CM is provided.The MAC of the single chip DOCSIS/EuroDOCSIS CM 1210 of the FIG. 12 isoperable to support all digital reference frequency lockingfunctionality according to the present invention. FIG. 12 shows yetanother embodiment where interference cancellation functionality may besupported according to the present invention. The interferencecancellation functionality may be viewed as being supported andperformed within the DOCSIS 2.0 MAC of the single chip DOCSIS/EuroDOCSISCM 1210 of the FIG. 12.

FIG. 13 is a system diagram illustrating an embodiment of a single chipwireless modem system 1300 that is built according to the presentinvention. The single chip wireless modem system 1300 includes a singlechip wireless modem 1310 that is operable to support a variety offunctionalities. The single chip wireless modem system 1300 is operableto perform wireless LAN operation using an 802.11 radio that is operableto communicatively couple to an external device that is wireless capable(example shown as the pen computer having wireless functionality). Thesingle chip wireless modem 1310 of the single chip wireless modem system1300 employs a 10/100 Ethernet PHY and an HPNA (Home Phoneline NetworkAlliance) analog front end (AFE) that is operable to interface with theHPNA 2.0 network. The single chip wireless modem 1310 of the single chipwireless modem system 1300 also supports capability to communicate withan external device via a USB 1.1 interface.

The single chip wireless modem 1310 of the single chip wireless modemsystem 1300 is compatible with existing cable modem application code. Inaddition, the single chip wireless modem 1310 supports advanced QAMLink®modulation/demodulation TP provide for higher throughputs andperformance in noisy plant environments. The 802.11b MAC and basebandallow for wireless connectivity as mentioned above. In addition, theintegrated HPNA 2.0 MAC supports high-speed multimedia services overphone lines. The integrated 10/100 Ethernet and USB 1.1 with integratedtransceiver provide for a low cost CPE (Customer Premises Equipment),and the MPI interfaces provide for great flexibility through additionalconnectivity options. The single chip wireless modem 1310 is a part of acomprehensive solution that is operable to support certifiableDOCSIS/EuroDOCSIS 1.1 software as well as supporting residential gatewaysoftware including Firewall, NAT, and DHCP. The FIG. 13 shows yetanother embodiment where interference cancellation functionality may besupported according to the present invention. The interferencecancellation functionality may be viewed as being supported andperformed within the single chip wireless modem 1310 of the single chipwireless modem system 1300.

FIG. 14 is a system diagram illustrating another embodiment of a singlechip wireless modem system 1400 that is built according to the presentinvention. The single chip wireless modem system 1400 includes a singlechip wireless modem that is operable to support a variety offunctionalities. The single chip wireless modem of the single chipwireless modem system 1300 integrates the DOCSIS/EuroDOCSIS 2.0 cablebased modem with a 2/4/16/32/64/128/256/1024 QAM downstream receiverwith Annex A, B, C FEC support. In addition, the single chip wirelessmodem integrates the DOCSIS/EuroDOCSIS 2.0 cable based modem with2/4/8/16/32/64/128/256 QAM FA-TDMA ad S-CDMA. The 802.11b wireless MACand baseband are also integrated on the single chip wireless modem. Anumber of other functional blocks are also integrated thereon,including, a 300 MHz MIPS32 CPU, a 32 bit 100 MHz SDRAM/DDR controller,an integrated upstream amplifier, an integrated IP SEC engine, anintegrated advance PROPANE™ packet accelerator, a 12 Mbps USB 1.1 slaveport with integrated transceiver, a 10/100 Ethernet MAC/PHY with MIIinterface, an MPI expansion bus (that supports PCI, Cardbus, and PCMCIAinterfaces), a single 28 MHz reference crystal, and ability to operateusing voltages of 1.8 V and/or 3.3 V.

The advanced QAMLink® technology of the single chip wireless modem,compliant with DOCSIS 2.0, supports up to 1024 QAM downstream modulationformats and both FA-TDMA and S-CDMA, with 256 QAM upstream modulationformats. This advanced technology provides a higher throughput andsuperior performance in noise plant environments, paving the way forsymmetrical services, such as video conferencing.

The single chip wireless modem integrates both wireless and wire linenetworking functions for distributing broadband content throughout thehome. An 802.11b solution is provided for wireless connectivity, whileboth 10/100 Ethernet and 32 Mbps HPNA 2.0 solutions provide wiredconnectivity. HPNA 2.0 allows multimedia services to be streamed acrossexisting home phone lines.

The PROPANE™ technology provides bandwidth and performance enhancementsto existing cable plants allowing up to twice as many subscribers pernode, thereby minimizing the need for node splits. The FIG. 14 shows yetanother embodiment where interference cancellation functionality may besupported according to the present invention. The interferencecancellation functionality may be viewed as being supported andperformed within the single chip wireless modem of the single chipwireless modem system 1400.

FIG. 15 is a diagram illustrating an embodiment of a vector de-spreader1500 that is built according to the present invention. The followingdescription of embodiments of the present invention using the FIGS. 15,16, 17, and 18 are made within the context of the DOCSIS 2.0 system.This system uses S-CDMA modulation for the upstream with 128 orthogonalcodes. In the example there are 120 active (data-carrying) codes, with 8unused (non data-carrying) codes. This example is for illustrativepurposes only, and should by no means limit the scope of the invention.Again, it is noted that the specific examples of 120 active codes, and 8unused codes in a system having 128 available codes is exemplary.Clearly, other embodiments may be employed (having different numbers ofcodes—both different numbers of used and unused) without departing fromthe scope and spirit of the invention.

The FIG. 15 depicts a vector de-spreader, arranged according to thepresent invention, consisting of 128 individual de-spreaders.De-spreading is the process of multiplying by a given code sequence andsumming (or integrating) over the chips of a spreading sequence, in thiscase the length of the code, 128 chips. Each scalar de-spreader performsthe function of de-spreading the received signal (input spread signal tobe de-spread) using a single de-spreading code (c₁, . . . , c₁₂₈). Thereare 128 orthogonal de-spreading codes in the present example.

FIG. 16 is a diagram illustrating an embodiment of an interferencecanceller 1200 that is built according to the present invention. Thespread input signal x, consisting of the sum of multiple spreading codesmodulating multiple data streams, enters the diagram at the left. Theundesired interference n is added to the signal. The signal is appliedto the vector de-spreader, which de-spreads each of the 128 codes. Theupper 8 codes are not used for data transmission and are modulated withnumerically zero-valued symbols instead of data. Clearly, there may beembodiments where other numerically constant-valued symbols orknown-value symbols may be employed instead of data as well. Further,the symbol may contain data represented as a reduced constellation, suchas BPSK or QPSK, on the “unused” codes.

One of the 120 data-carrying codes, code d_(s), is identified forillustration in the FIG. 16. In order to cancel the interference, thede-spreader output d_(s) is processed in a linear combiner, where it issummed with a linear combination of the 8 de-spreader outputs from thezero-modulation codes d₁-d₈. The complex-valued combining weightsapplied to these codes are w₁-w₈, respectively. These weights arecomputed in a weight computation method as shown in the lower right handcorner of the FIG. 16 using the weight computation functionality.

The weight computation functionality may employ a method that utilizesthe input spread signal plus interference, and may utilize some systemoutputs if an iterative method is used. Examples of weight computationmethods that have been found valuable are the LMS (least mean square)method and the LS (least squares) method. The result of the linearcombination is the output {circumflex over (d)}_(s), which is the datastream d_(s) with the interference largely removed. Although not shownin the figure, the same linear combiner structure is applied to theother 119 codes as well (all of the other active codes besides the coded_(s)). In each case, the desired code (one of the 120 “active” ordata-carrying codes) is applied to a linear combiner to cancel theinterference from that code. For each data-carrying code, the same 8zero-modulation codes are summed with the desired code, but for eachactive code the weights w₁-w₈ are in general unique.

An alternative viewpoint is to define the adapted code as

${c_{a}(n)} = {{c_{s}(n)} + {\sum\limits_{k = 1}^{N_{u}}\;{w_{k}{c_{k}(n)}}}}$that is, the desired code plus the linear combination of the weightstimes the unused (“inactive”) codes. In this view, the adapted code is amodified code with complex coefficients, which is used instead of thecode c_(s) to de-spread a single desired signal from a single modulatedcode, while simultaneously canceling the interference. A weight couldalso be applied to the desired signal, but usually this weight is 1.

This approach can be extended to matrix notation by defining, theadapted code matrix asC _(adapted) =C _(used) +WC _(unused)where:

C_(adapted)=adapted code matrix, dimension (N_(c)-N_(u))×128, forexample, 120×128

C_(used)=matrix whose rows are the used codes in the original codematrix, dimension (N_(c)-N_(u))×N_(c), for example, 120×128

W=matrix whose rows are the adaptive weight vectors for each unusedcode, dimension (N_(c)-N_(u))×N_(u), for example, 120×8

C_(unused)=matrix whose rows are the unused codes in the original codematrix, dimension N_(u)×N_(c), for example, 8×128

N_(c)=number of total codes=number of chips in each code, for example,128

N_(u)=number of unused codes, for example, 8

In this view, the adapted code matrix is a modified code matrix withcomplex coefficients, which is used instead of the code matrix C tode-spread the desired signals from all used codes, while simultaneouslycanceling the interference on all used codes.

It is noted that that the unused codes may be de-spread as well, andthis side information, though not data-carrying, is of use incharacterizing the interference environment.

It is also noted that the weight computation functionality may beperformed offline, and these pre-computed complex-valued combiningweights, w₁-w₈, may then be stored in memory and/or a look up table(LUT) that may be used to provide the complex-valued combining weights,w₁-w₈. The appropriate set of weights may be selected after analyzingthe interference environment.

FIG. 17 is a diagram illustrating another embodiment of an interferencecanceller 1700 that is built according to the present invention. TheFIG. 17 may be viewed as being somewhat similar to the interferencecanceller 1600 of the FIG. 16 with some exceptions relating to thespecific codes that are used to perform the linear combination in aneffort to perform the interference cancellation according to the presentinvention.

FIG. 17 shows an embodiment where all of the codes are included in thelinear combiner. This includes both the used codes and the unused codes,instead of only the unused codes. This will be useful if there isinter-code interference (ICI), since in that case the desired signalwill appear on all codes. Conversely, the signals modulated onto allcodes will appear on the desired de-spreader output, and can besubtracted from the desired de-spreader output.

In yet another alternative embodiment, to add to the number of effectiveunused codes, we may use codes bearing preamble symbols in addition tothe codes carrying zero-valued symbols. The preamble symbols are knownand can be subtracted once their amplitude and phase have been measured,for example using a preamble correlator. Thus the preamble-bearing codescan also be used as inputs to the linear combiner in order to bettercancel the interference.

There are some other embodiments that may be employed as well. Forexample, the selection of the inactive codes may be performed asfollows: (1) use codes 0, 1, 2, 3, . . . n (adjacent codes, as done inDOCSIS 2.0 spec) in which the codes are adjacent and the lower codesused in the coding and/or (2) spacing the codes maximally apart. Forexample, using DOCSIS 2.0 S-CDMA code set, the 8 unused codes out of 128total codes might be code numbers {15 31 47 63 79 95 111 127} whenseeking to perform the maximally spaced apart embodiment. Moreover, theselection of the unused codes may be performed according to anoptimality criterion. Examples of some potential optimality criteriainclude: (1) select unused codes that have maximal correlation with theinterference, (2) minimize enhancement of white noise resulting fromcancellation process, and (3) minimize residual interference power aftercancellation.

It is also noted that the particular codes that are selected as theunused codes may change over time during the processing of receivedsignals. Moreover, the particular selection of the codes may vary fromone iteration to the next. For example, in one situation, adjacent codesmay be selected as the unused codes. In another situation, the maximallyspaced codes may be selected as the unused codes.

The selection of the codes that are to be designated the unused codesmay be performed using a variety of approaches including: (1) employingcode matrix reordering, (2) employing mill grant periods, (2) zeropadding data, and/or (4) employing some optimality criterion (orcriteria).

Similar to the embodiment of the FIG. 16, it is also noted that theweight computation functionality may be performed offline, and thesepre-computed complex-valued combining weights that are used here in theFIG. 17 may similarly be stored in memory and/or a look up table (LUT)that may be used to provide the complex-valued combining weights. Theappropriate set of weights may be selected after analyzing theinterference environment. This may similarly be performed in theembodiments of the FIGS. 18 and 19 described in further detail belowwhere these pre-computed complex-valued combining weights may also bestored in memory and/or a look up table (LUT).

FIG. 18 is a diagram illustrating another embodiment of an interferencecanceller 1800 that is built according to the present invention. TheFIG. 18 may be viewed as being a variant of the FIG. 17 that has accessto any of the codes (including both used and unused codes). The FIG. 18includes a subset of the codes for use in the linear combiner, insteadof all the codes or only the unused codes. For example, in DOCSIS 2.0S-CDMA, adjacent codes are nearly shifts of each other. When a timingoffset occurs, the codes lose orthogonality and ICI occurs. However, theICI is predominant on the adjacent codes. For example, in the presenceof a timing offset, code 35 will be interfered with predominantly bycodes 34 and 36, with lesser effects coming from codes 33 and 37, evenlesser effects from codes 32 and 38, and so on. Hence including asinputs to the linear combiner the data-bearing codes 33, 34, 36 and 37,plus the unused codes, but not the remaining data-bearing codes, willreduce the number of weights that have to be solved for compared to themore general case above in which all codes are included in the linearcombination. Another embodiment would involve including as inputs to thelinear combiner the data-bearing codes 34, and 36, plus the unusedcodes, but not the remaining data-bearing codes, in an effort to try toreduce the number of weights that have to be solved for compared to themore general case above in which all codes are included in the linearcombination.

FIG. 19 is a diagram illustrating an embodiment of an interferencecanceller with memory 1900 that is built according to the presentinvention. The FIG. 19 shows an interference canceller with memory, thatis, it uses the history of previous samples in computing the output. Theweight w₁ has been replaced with “feed-forward equalizer 1” (FF Eqer.1), a tapped delay line or FIR filter with L weights. The other adaptiveweights have similarly been replaced with FF equalizers 2-8. It is notedthat both the current and past soft de-spread symbols are included inthe linear combination. Moreover, future soft de-spread symbols may alsobe included in the linear combination; these future soft de-spreadsymbols are “future” relative to the symbol currently being estimated.This permits each tap to have a frequency selective response.

FIG. 20 is a diagram illustrating an embodiment of equalization withcanceller 2000 that is arranged according to the present invention. TheFIG. 20 may be viewed as being a representation of equalization ofchannel response, and cancellation of resulting colored noise. Thecanceller structure can also be used to help with equalization. The FIG.20 considers a communications system with a transmitter 2010, a channel2020 having a response H(f), and a receiver 2025. Let the channelresponse be H(f). We assume, as an example, that H(f) exhibits a null atsome frequency in the signaling band of interest. Assume AWGN (additivewhite Gaussian noise) is added in the channel after H(f). We may use astandard adaptive equalizer 2030 at the receiver to provide the inverse(zero forcing) response, 1/H(f), which will have a narrow peak at thefrequency location where the null exists in H(f). This peak will causethe white noise to be colored and to have a peak as well. Thisnarrowband colored noise can be canceled (using the canceller 2040) bythe present technique in exactly the same manner that other narrowbandinterference is canceled. The FIG. 20 shows yet another embodiment ofhow interference cancellation, according to the present invention, maybe performed.

FIG. 21 is a diagram illustrating an embodiment of Least Means Square(LMS) training of an interference canceller 2100 according to thepresent invention. The FIG. 21 may be viewed as being one embodimentthat is operable to perform adaptation of an interference cancellerusing iterative methods. The present interference canceller 2100 can beadapted using iterative methods such as LMS or RLS. The FIG. 21illustrates how the LMS method may be used to adapt the cancellerweights. The output of the de-spreader for the desired code (containingsoft decisions) is sliced to produce hard symbol decisions. If knowntraining symbols are available, they replace the hard decisions, whichmay contain symbol errors, especially upon startup. The differencebetween the hard (or known) and soft decisions gives the LMS errorsample. The error is correlated with the outputs of the unused codede-spreaders and used to update the adaptive weights w_(i).

Within the FIG. 21, the slicer, the MUX, and the LMS error (and LMSstep-size scaling μ) that are used to update the adaptive weights w_(i)may be viewed as being just one embodiment of an iterative, errordetermining approach. Clearly, other error determining approaches(besides LMS) may be employed without departing from the scope andspirit of the invention. The error calculation and correlation with theoutputs of the unused code de-spreaders that are used to update theadaptive weights w_(i) may be viewed as being an iterative adaptiveweight functionality that may be viewed as being provided in animplementation via an iterative adaptive weight functional block thatcommunicatively couples to each of the outputs of the unused codede-spreaders.

FIG. 22A is a diagram illustrating an embodiment of signaltransformation according to the present invention. The FIG. 22A includesthe pre-processing of an input signal, and unused inputs, via anorthogonal transformation 2210 to generate a representation of the inputsignal within a finite signal space. The orthogonal transformation 2210may be an orthonormal transformation in certain embodiments. Now thatthe input signal is represented in the finite signal space, the signalis then passed through a communication channel 2230 after which it isprovided to an interference cancellation functional block 2220 that isoperable to perform any of the various embodiments of interferencecancellation described herein. The communication channel 2230 mayintroduce interference. It is noted that the present invention isoperable to perform cancellation of interference of a variety of typesincluding (1) narrowband interference in general, (2) Ham radio, CBradio and HF radio, (3) adjacent channel interference (spillover fromdesired signals in neighboring channels), (4) CDMA on a small number ofcodes, and (5) impulse/burst note. The present invention envisions anyorthogonal transformation 2210 that is operable to transform an inputsignal into a representation of a finite number of elements within afinite signal space so as to facilitate the interference cancellationaccording to the present invention.

FIG. 22B is a diagram illustrating another embodiment of signaltransformation according to the present invention. The FIG. 22B includesthe pre-processing of an input signal, and unused inputs, via anidentity matrix transformation 2215 to generate a representation of theinput signal within a finite signal space. Again, the orthogonaltransformation 2215 may be an orthonormal transformation in certainembodiments. Now that the input signal is represented in the finitesignal space, the signal is then passed through a communication channel2235 after which it is provided to an interference cancellationfunctional block 2225 that is operable to perform any of the variousembodiments of interference cancellation described herein. Thecommunication channel 2235 may introduce interference. It is again notedthat the present invention is operable to perform cancellation ofinterference of a variety of types including (1) narrowband interferencein general, (2) Ham radio, CB radio and HF radio, (3) adjacent channelinterference (spillover from desired signals in neighboring channels),(4) CDMA on a small number of codes, and (5) impulse/burst note. Furtherdetails are described below with respect to impulse noise cancellation.

Impulse noise is nearly zero most of the time, and large during a fewsamples. For the purpose of analysis only, we consider the rows of theN×N identity matrix as the basis set, where N is the number of samplesper frame (or chips per spreading interval) under consideration. In thisbasis set, each time sample represents one dimension. Hence we see thatthe impulse noise only occupies a small number of dimensions. Thus itcan be canceled by this technique, using an arbitrary basis set, such asthe S-CDMA codes. The adapted de-spreading code has zeros (or nearlyzeros) at the chips corresponding to the time location of the impulsenoise. However, impulse noise occurs at a random, unpredictable locationin each frame. If we know where it is, we can solve the equations forthe weights. But the next frame it will be in a different place. Thismeans re-doing the computations every frame, resulting in highcomplexity.

For low-level impulse noise, it may be difficult to locate the chipsthat are affected by impulse noise. We may use one or more “indicatorcodes” for this purpose, as follows. As an example, say we have 128total codes—for example in the DOCSIS 2.0 situation. We designate 119 ofthese codes as used, or data-carrying codes. We designate 9 of the codesas unused codes, on which numerically-zero-valued symbols aretransmitted. Of these 9 unused codes, 8 codes participate in the linearcombiner for noise cancellation, and there is 1 extra or “indicator”code. The indicator code is de-spread as if it were a used code, thatis, it is given the benefit of the linear combiner canceller. We expectto get a zero symbol at its de-spreader output; if we see noise insteadof zero that provides an indication of the amount of noise that has notbeen canceled. We then proceed as follows to locate the impulse noise.Assume for example that there is one occurrence of impulse noise in agiven symbol, and that the impulse noise affects 8 or fewer chips. Webegin with a set of weights w that null chips 1 through 8 in the timedomain. We use w to de-spread the indicator code, and observe the outputy. We then modify w to null chips 2-9, and again observe y. In a similarmanner, we scan w across the entire symbol, measuring y at each timeoffset. The power |Y|² will exhibit a minimum for the weight set w thatcorresponds to the time location of the impulse noise. In this mannerthe location of the impulse noise can be determined. Once located, itcan be canceled.

FIG. 23 is an operational flow diagram illustrating an embodiment of aninterference cancellation method 2300 that is performed according to thepresent invention. In a block 2310, a spread signal is received thatcontains interference. Then, the received spread signal is de-spreadinto a number of codes in a block 2320. Each of the codes is selectivelyprocessed using linear combination processing as shown in a block 2330.There are a variety of ways in which the linear combination processingmay be performed according to the present invention including using anumber of unused codes, using all of the available codes, and/or usingselected adjacent codes in addition to the unused codes. Ultimately, theinterference cancelled de-spread codes are output as shown in a block2340.

FIG. 24 is an operational flow diagram illustrating another embodimentof an interference cancellation method 2400 that is performed accordingto the present invention. Initially, in some embodiments, the methodinvolves selecting those codes that are to be used as the unused codesas shown in a block 2402. As shown within the FIG. 24, there are threedifferent ways in which this may be performed. They include code matrixreordering, employing null grant periods, and/or zero padding data. Evenother ways are described when referring to the other Figures as well.These will be the codes that are used to perform the linear combining toeffectuate the interference cancellation according to the presentinvention. In even other embodiments as shown in a block 2404, theunused codes (N_(u)) are modulated with numerically zero-valued symbols.Alternatively, the unused codes (N_(u)) may be modulated withnumerically constant-valued symbols that are non-zero or withknown-value symbols without departing from the scope and spirit of theinvention.

In a block 2410, a spread signal is received that contains interference.Then, the received spread signal is de-spread into a number of codes(N_(c)) as shown in a block 2420. in a block 2430, each of the number ofunused codes (N_(u)) is selectively de-spread. The method then willcontinue to the block 2440 in most instances.

However, in certain embodiments, the method will continue from the block2430 to the block 2432 in which each of the number of preamble codes isselectively de-spread. The preamble symbols are known and can besubtracted once their amplitude and phase have been measured, as shownin a block 2434, for example using a preamble correlator. Thus thepreamble-bearing codes can also be used as inputs to the linear combinerin order to better cancel the interference.

As shown in the block 2440, complex-valued weights for linearcombination processing of the unused codes (N_(u)) are selectivelycalculated. This processing in the block 2440 may be performed byinputting the spread signal, interference, and/or outputs as shown in ablock 2442. In the embodiments where the blocks 2432 and 2434 areperformed, the preamble-bearing codes may be input as shown in a block2444 when performing the processing in the block 2440. The processing inthe block 2440 may be performed be employing LMS processing as shown ina block 2446 and/or LS processing as shown in a block 2448.

Then, in a block 2450, the complex value weights are selectively appliedto scale the unused codes (N_(u)). In a block 2460, the now scaledunused codes (N_(u)) are selectively summed with the desired code.Ultimately, the interference cancelled de-spread codes are output asshown in a block 2440.

FIG. 25 is an operational flow diagram illustrating an embodiment of anunused code selection method 2500 that is performed according to thepresent invention. The question arises whether any subset of the codesis a good choice for the unused codes. We consider the example ofnarrowband interference cancellation in a DOCSIS 2.0 S-CDMA system. Forefficient narrowband interference canceling capability, the unused codeshave to be chosen such that it is possible to combine them in theadapted codes (in the linear combiner) to form one or more notches inthe frequency domain. Thus, for optimal performance, one might need todesignate specific codes as unused. The current DOCSIS 2.0 draftspecification does not permit the selection of which codes are unused.It has been found that successive, or “adjacent,” DOCSIS 2.0 codes arenot a good choice. This is because each code is approximately a shift ofthe previous code. This implies that adjacent or nearly adjacent codeshave nearly the same frequency response. Some techniques that could beused to “force” unused codes at specific rows of the code matrix are thefollowing:

The code matrix may be reordered. In this technique, both the CM andCMTS re-order the code matrix as shown in a block 2510, prior tospreading or de-spreading. This may be performed such that desiredunused codes are grouped together (say at the lower part of therearranged code matrix) as shown in a block 2520. Similarly, the desiredused codes should be grouped as well (say at the upper part of therearranged code matrix) as shown in a block 2530. Using such techniquerequires the knowledge of the reordering pattern at the CMTS as well asall CMs; this may be ensured as shown in a block 2540.

Alternatively, the selection of unused codes may be performed using nullgrant periods. In this technique, the CMTS instructs all CMs to besilent during a specific grant as shown in a block 2505 (i.e., thedesired unused codes). This technique has the advantage that the CMsneed not have prior knowledge of the unused codes and just follow theCMTS grants. However, it may be viewed as causing inefficiencies to theCMTS scheduling process that may prohibit this approach in someimplementations.

Alternatively, the selection of unused codes may be performed byzero-padding the data. In this technique, the CMTS grants the CM alonger grant period that what is needed to transmit the grant data asshown in a block 2555. If desired when performing the operation of theblock 2555, the grant sizes are chosen by the CMTS in a way such thatthe CM zero-padding occurs at the desired unused codes as shown in ablock 2557. The CMTS also instructs the CM to append the transmitteddata with zero-symbols as shown in a block 2565.

FIG. 26 is an operational flow diagram illustrating an embodiment of anS-CDMA interference cancellation method 2600 that is performed accordingto the present invention. In a block 2610, a set of used codes isselected. Then, in a block 2620, a set of unused codes is selected. Asignal is transmitted using the used codes as shown in a block 2630. Incertain embodiments, as shown in a block 2632, we transmit one or morezeroes on the inactive/unused codes. Alternatively, as shown in a block2634, we transmit a known sequence (training or pilot symbols) on theinactive/unused codes. Alternatively, we may transmit lower ordermodulation on “inactive” codes.

Then, in a block 2640, the received signal is processed using thereceived signal's projection on the active (used) codes and the inactive(unused) codes thereby canceling interference. We process the receivedsignal using both its projection onto a desired (active) code, and itsprojection onto the inactive codes, in order to cancel interference onthe desired code. From certain perspectives, in the context of a systemand method that employ vector de-spreading to a spread signal, theprojection may be viewed as being the vector de-spreader output.However, in other contexts, a projection may be viewed as being therepresentation of the received signal across its finite signal space.This understanding of projection may be used to describe therepresentation of the signal across a finite signal space.

The operation of FIG. 26 is performed within the context of an S-CDMAcommunications system in the presence of interference. There are anumber of types of S-CDMA systems that may support the method of theFIG. 26. For example, some types of S-CDMA systems include DOCSIS 2.0set of codes and Walsh-Hadamard codes. The selection of theinactive/unused codes may be performed as illustrated and describedabove within some of the embodiments shown in a block 2621 including usecodes 0, 1, 2, 3, . . . (adjacent codes, as done in DOCSIS 2.0 spec),spacing the inactive/unused codes maximally apart as shown in a block2622, and selecting the codes according to an optimality criterion asshown in a block 2623.

For example, when spacing the inactive/unused codes maximally apartwithin DOCSIS 2.0 S-CDMA code set, the 8 unused codes out of 128 totalcodes might be code numbers {15 31 47 63 79 95 111 127}. When selectinga different number of unused codes within the DOCSIS 2.0 S-CDMA codeset, they may be similarly maximally spaced apart.

In addition, the inactive/unused codes may be selected according to theoptimality criterion of the block 2623. Examples of an optimalitycriteria would be to select unused codes include: (1) selectinginactive/unused codes that have maximal correlation with theinterference as shown in a block 2624, (2) minimizing enhancement ofwhite noise resulting from cancellation process as shown in a block2625, and (3) minimizing residual interference power after cancellationas shown in a block 2626.

FIG. 27 is an operational flow diagram illustrating another embodimentof an interference cancellation method 2700 that is performed accordingto the present invention. In a block 2710, a set of active/used basiswaveforms is selected. These basis waveforms may include orthogonal (ornearly orthogonal) waveforms; these waveforms may be viewed as beingsubstantially orthogonal. There are a number of types of sets oforthogonal waveforms may be employed. Some specific examples of sets oforthogonal waveforms include: (1) S-CDMA codes, including DOCSIS 2.0 andWalsh-Hadamard, (2) an orthogonal set of binary spreading codes, (3) anyorthogonal set of quaternary spreading codes, and (4) the rows of theidentity matrix.

Then, in a block 2720, a set of inactive/unused basis waveforms isselected. In certain embodiments as shown in a block 2722, we may assumethat the number of inactive/unused basis waveforms is less than numberof active/used basis waveforms. A signal is transmitted using theactive/used basis waveform is as shown in a block 2730. In certainembodiments, as shown in a block 2732, we transmit one or more zerovalued symbols on the inactive/unused basis waveforms. Alternatively, asshown in a block 2734, we transmit a known sequence (training or pilotsymbols) on the inactive/unused basis waveforms.

Then, in a block 2740, the received signal is processed using thereceived signal's projection on the active (used) basis waveforms andthe inactive (unused) basis waveforms thereby canceling interference. Weprocess the received signal using both its projection onto a desired(active) waveform, and its projection onto the inactive waveforms, inorder to cancel interference on the desired waveform.

Alternatively, in a block 2742, the received signal is processed usingthe received signal's projection on the active (used) basis waveformsthereby canceling interference. We process the received signal using itsprojection onto a desired (active) waveform in order to cancelinterference on the desired waveform.

In even alternative embodiments, in a block 2744, we compute theprojection of the interference on the inactive/unused basis waveforms,and subtract it from the projection on the active/used basis waveformsof the received signal including interference thereby cancelinginterference. It is noted here that we can reduce the computationalcomplexity by computing the null-space projection (the projection of theinterference on the inactive basis waveforms) and subtracting it fromthe overall projection (the projection of the signal + interference onthe active basis waveforms). As an example, if we have 120 active codesand 8 inactive codes, in the present method we only need to invert an8×8 matrix. In the standard least-squares approach, we would have toinvert a 120×120 matrix.

The selection of the inactive/unused basis waveforms may be performed asillustrated and described above within some of the embodiments selectingadjacent basis waveforms, spacing the inactive/unused basis waveformsmaximally apart, and selecting the basis waveforms according to anoptimality criterion. Examples of optimality criteria would be to selectunused basis waveforms include: (1) selecting inactive/unused basiswaveforms that have maximal correlation with the interference, (2)minimizing enhancement of white noise resulting from cancellationprocess, and (3) minimizing residual interference power aftercancellation. These parameters that may be used to perform the selectionof the inactive/unused basis waveforms is analogous to the selection ofthe inactive/unused codes that is performed above with respect to theFIG. 26, except here, the selection is with respect to the basiswaveforms of the signal space.

FIG. 28 is a diagram illustrating an embodiment of a spectrum ofnarrowband interference 2600 that may be addressed and overcome whenpracticing the present invention. The FIG. 28 shows the spectrum ofnarrowband interference (for example, the signal n in the FIGS. 16, 17,and/or 18) that may be present at the input of a communicationsreceiver. The desired signal is not present in this Figure. In thisexample, the 3-dB bandwidth of the interference is 1/32 of the symbolrate of the desired signal. Its power is equal to the desired signal (0dBc), when the desired signal is present. The SNR (Signal to Noise) ofthe desired signal, when present, is 35 dB in the example.

FIG. 29 is a diagram illustrating an embodiment of a spectrum of anadapted code showing null at a location of interference 2900 that may beachieved when practicing via the present invention. The FIG. 29 showsthe spectrum of the adapted code. The adapted code is seen to have anull corresponding to the narrowband interference. Hence, the adaptedcode cancels the narrowband interference.

FIG. 30A is a diagram illustrating an embodiment of a receivedconstellation before interference has been cancelled 3000 whenpracticing the present invention. The FIG. 30A shows the output d_(s) ofthe vector de-spreader before interference cancellation is enabled usingthe linear combiner and weight computation functionality. This may beviewed as being the output d_(s) within any of the FIGS. 16, 17, and/or18. That is, all the adaptive weights w_(i) are zero, and only thenormal de-spreading code c_(s) is used to de-spread the desired signal.We see that the signal constellation is unrecognizable due to the largeamount of interference, which has not yet been canceled.

FIG. 30B a diagram illustrating an embodiment of a receivedconstellation after interference has been cancelled 3005 when practicingvia the present invention. The FIG. 30B shows the output d_(s) of thede-spreader after the interference cancellation of the present inventionis enabled using the linear combiner and weight computationfunctionality. Now the adaptive weights w_(i) have adapted and arenonzero, and as a result the adapted de-spreading code c_(a) is used tode-spread the desired signal. We see that the 64 QAM signalconstellation, plus the QPSK constellation used for the preamble, is nowclearly recognizable and the interference has been effectively canceled.

FIG. 31 is a flow chart illustrating an embodiment of a method of thepresent invention for removing burst noise from a received spreadsignal. With this method, a communication receiver receives a spreadsignal that was formed at a transmitter from at least one known-valuesymbol spread by a plurality of non data-carrying orthogonal codes anddata symbols spread by at least one data-carrying orthogonal code. Anyknown set of symbols spread by the non-data-carrying codes may beemployed. These known-value symbols can be constant or varying in somedetermined manner, as long as they are known to the receiver. Numericalzeros modulating the non-data-carrying codes are one of the simplestcases to implement. This spread signal was transmitted on a channel thatintroduced burst noise and was received with the burst noise (step3102). After receipt, the receiver determines chips of the orthogonalcodes that are affected by the burst noise (step 3104). One techniquefor determining the chips that are affected by the burst noise isdescribed further in U.S. utility patent application Ser. No.10/962,803, filed Oct. 12, 2004, now U.S. Pat. No. 7,366,258, issued onApr. 29, 2008, and entitled, “CHIP BLANKING AND PROCESSING IN SCDMA TOMITIGATE IMPULSE AND BURST NOISE AND/OR DISTORTION,” and having commoninventorship and common ownership with the present application. Themethod continues with calculating a plurality of complex-valuedcombining weights based upon the burst noise affected chips (step 3106).One technique for determining such complex-valued weights for burstnoise interference will be described further herein after description ofFIG. 38. However, other techniques can of course be used, some of whichhave been previously described with reference to FIGS. 22-27. The methodconcludes by using the plurality of non data-carrying orthogonal codes,the at least one data-carrying orthogonal code, and the plurality ofcomplex-valued combining weights to de-spread the received spread signalto produce the data symbols with the burst noise substantially removed(step 3108). Various embodiments of the operations of step 3108 aredescribed further herein with reference to FIGS. 33, 34, 36, and 37.Each of these various embodiments of the preset invention supportmultiple modulation types, including binary modulation types, quadraturemodulation types, and higher order modulation types such as QAMmodulations.

FIG. 32 is a graph illustrating the manner in which burst noise mayaffect a portion of the chips of a symbol of the spread signal,referenced with respect to chips of an orthogonal code, e.g., the signaln in FIGS. 16, 17, and/or 18 that may be present at the input of acommunications receiver. In the example of FIG. 32, burst noise ispresent during chips 94-100 of a 128 chip sequence S-CDMA code. Using adifferent approach for interference suppression, one could simply startwith the normal S-CDMA code c_(s) and arbitrarily replace a group ofchips (those numbered 94-100 in the example) with zeros, or with verysmall values. However, such a modified code would no longer beorthogonal to the remaining S-CDMA codes. This loss of orthogonalitywould result in inter-code interference (ICI). While the ICI could beremoved by a successive interference canceller (SIC) as described inco-pending application Ser. No. 10/242,032, filed on Sep. 12, 2002,entitled SUCCESSIVE INTERFERENCE CANCELLATION FOR CDMA, and havingcommon inventorship and common ownership with the present application,such solution results in reduced performance in many cases compared tothe present invention. In contrast, the communication system of thepresent invention complexly combines at least the non data-carryingcodes so that the combination of adjusted codes causes the burst noiseaffected chips (94-100 in the example) to have vanishingly small valueswhile maintaining the orthogonality of the spreading codes that are usedto distinguish signals of interest.

FIG. 33 is a flow chart illustrating a first embodiment of the method ofFIG. 32 that may be implemented with the structure of FIG. 16. With theembodiment of FIG. 33, step 3108 of FIG. 31 includes despreading thespread signal using the plurality of non data-carrying orthogonal codesto produce a plurality of non data-carrying despread signals (step3302). The plurality of non data-carrying orthogonal codes are referredto as c₁, c₂, . . . c₈ in FIG. 16. The non data-carrying despreadsignals are referred to as d₁, d₂, . . . d₈ of FIG. 16. Step 3108further includes despreading the spread signal using the at least onedata-carrying orthogonal code to produce at least one data-carryingdespread signal (step 3304), i.e., d_(s) of FIG. 16. With these stepscompleted, the method includes applying the plurality of complex-valuedcombining weights (w_(i)-w_(s) of FIG. 16) to the plurality of nondata-carrying despread signals to produce a plurality of complex-valuedadjusted non data-carrying despread signals (step 3306). The operationsof step 3108 finally include combining the plurality of complex-valuedadjusted non data-carrying despread signals with the at least onedata-carrying despread signal to produce the data symbols with the burstnoise substantially removed (step 3308).

FIG. 34 is a flow chart illustrating a second embodiment of the methodof FIG. 32 that may be implemented with the structure of FIG. 17. Withthe embodiment of FIG. 34, step 3108 of FIG. 31 includes despreading thespread signal using the plurality of non data-carrying orthogonal codesto produce a plurality of non data-carrying despread signals (step3402). The plurality of non data-carrying orthogonal codes are referredto as c₁, c₂, . . . c₈ in FIG. 17. Step 3108 further includesdespreading the spread signal using the at least one data-carryingorthogonal code to produce at least one data-carrying despread signal(step 3404). The at least one data-carrying orthogonal code referred toas c_(s) in FIG. 17. Step 3108 then includes applying the plurality ofcomplex-valued combining weights (w₁-w₈ and w_(s) of FIG. 17) to theplurality of non data-carrying despread signals and to the at least onedata-carrying despread signal to produce a plurality of complex-valuedadjusted despread signals (step 3406). Step 3108 concludes withcombining the plurality of complex-valued adjusted despread signals toproduce the data symbols with the burst noise substantially removed(step 3408).

FIG. 35 is a diagram illustrating an embodiment of an interferencecanceller 3500 that is built according to the present invention. With afirst example of the interference canceller 3500, the linear combiner3502 is operable to: (1) apply the plurality of complex-valued combiningweights to the plurality of non data-carrying orthogonal codes toproduce a plurality of complex-valued weighted non-data carryingorthogonal codes; and (2) combine the plurality of complex-valuedweighted non-data carrying orthogonal codes with the at least onedata-carrying orthogonal code to produce an adapted rotated code. Insuch case, the weight w₁ is unity while at least some of weights w₂through w₈ are non-unity and complex-valued. With this example, thevector de-spreader 3504 is operable to despread the received spreadsignal using the adapted rotated code to produce the data symbols withthe burst noise substantially removed. A burst noise detector 3506 isoperable to detect chips of a code that are affected by burst noise.Weight computation block 3508 is operable to determine the weightsw₁-w₈.

With a second example of the interference canceller 3500, the linearcombiner 3502 is operable to: (1) apply the plurality of complex-valuedcombining weights to the plurality of non data-carrying orthogonal codesand to the at least one data-carrying orthogonal code to produce aplurality of complex-valued weighted orthogonal codes; and (2) combinethe plurality of complex-valued weighted orthogonal codes to produce anadapted rotated code. In such case, all of weights w₁ through w₈ may benon-unity and complex-valued. With this example, the vector de-spreader3504 is also operable to despread the received spread signal using theadapted rotated code to produce the data symbols with the burst noisesubstantially removed.

FIG. 36 is a flow chart illustrating a third embodiment of the method ofFIG. 32 with the structure of FIG. 35. Operation of step 3108 commenceswith applying the plurality of complex-valued combining weights to theplurality of non data-carrying orthogonal codes to produce a pluralityof complex-valued weighted non-data carrying orthogonal codes (step3602). Operation of step 3108 continues with combining the plurality ofcomplex-valued weighted non-data carrying orthogonal codes with the atleast one data-carrying orthogonal code to produce an adapted rotatedcode (step 3604). Operation of step 3108 concludes with despreading thespread signal using the adapted rotated code to produce the data symbolswith the burst noise substantially removed (step 3606). This embodimentof step 3108 of FIG. 31 corresponds to the first example described abovewith reference to FIG. 35.

FIG. 37 is a flow chart illustrating a fourth embodiment of the methodof FIG. 32 with the structure of FIG. 35. Operation of step 3108commences with applying the plurality of complex-valued combiningweights to the plurality of non data-carrying orthogonal codes and tothe at least one data-carrying orthogonal code to produce a plurality ofcomplex-valued weighted orthogonal codes (step 3702). Operation of step3108 continues with combining the plurality of complex-valued weightedorthogonal codes to produce an adapted rotated code (step 3704).Operation of step 3108 concludes with despreading the spread signalusing the adapted rotated code to produce the data symbols with theburst noise substantially removed (step 3706). This embodiment of step3108 of FIG. 31 corresponds to the second example described above withreference to FIG. 35.

FIG. 38 is a chart illustrating the manner in which a plurality ofrotated codes are employed to attenuate the burst noise illustrated inFIG. 31. The resulting adapted rotated code is nearly zero at theselected chips, but is still orthogonal to the other codes; that is, noICI is added. The example of FIG. 38 corresponds most closely with theinterference canceller of FIG. 35 and the operations of FIGS. 36 and 37.However, the principles described with regard to the other FIGs. arecorrectly related in FIG. 38, i.e., that the incoming signal isessentially zeroed for the burst noise affected chips without causingISI.

One technique for determining complex-valued weights is next described.The example described first is directly applicable to the structure ofFIG. 35. First, consider an S-CDMA system described by the equationr=Cx+vWhere

-   -   r is a length N column vector representing received signal plus        noise samples, one sample per chip.    -   x is a length N column vector [x₁ x₂ x₃ . . . x_(k) 0 0 0 . . .        0]^(T) representing N−k zeros followed by k transmitted data        symbols.    -   C is an N×N orthonormal code matrix whose columns are the N        orthonormal S-CDMA codes.    -   v is a length N column vector representing added burst noise        samples, one sample per chip.

In this formula, the vector x contains k data symbols that are to betransmitted through the channel. Each symbol modulates one code vectorin the code matrix C. In addition, there are N−k zero-valued symbols,which modulate N−k unused codes. These unused codes provide theredundancy that permits cancellation of interference in accordance withthe present invention. Multiplication of x by the code matrix Crepresents the spreading operation that occurs in the transmitter (ortransmitters). In a multiple-access system, this matrix multiplication(spreading) may be distributed across several transmitters, where eachsynchronized transmitter is assigned a subset of the codes. The spreadsignals from the various transmitters are added together in the channelin such a multiple-access system. The noise samples v represent addedburst noise that occurs in the channel.

In the despreading process, the receiver multiplies the received vectorr by the conjugate transpose C* of the code matrix. The despreaderoutput y is given byy=C*ry=C*Cx+C*vy=x+C*vWhere

-   -   y is a length N column vector representing despread signal plus        noise samples (soft decisions at despreader output).

Note that C*C=I, the identity matrix, since the codes are orthonormal.We are left with the original data vector x plus a despread noise vectorC*v. The code matrix can be partitioned into the used-code(data-carrying code) matrix C₁ and the unused-code (non data-carryingcode) matrix C₂:C=[C ₁ |C ₂]Where

-   -   C₁ is the N×k matrix containing the used codes, that is, the        matrix C with the columns corresponding to the unused codes        removed    -   C₂ is the N×(N−k) matrix containing the unused codes, that is,        the matrix C with the columns corresponding to the used codes        removed        Consider the N−k unused codes only. The despreader output is        given by        y ₂ =x ₂ +C* ₂ v        y ₂ =C* ₂ v        Where the subscript “2” indicates that only the N−k rows        corresponding to the unused codes are included in the matrices        and vectors:    -   y₂ is a length N−k column vector representing despread signal        plus noise samples (soft decisions at despreader output).    -   x₂ is a length N−k column vector [0 0 0 . . . 0]^(T)        representing the N−k zero symbols that are transmitted on the        unused codes.    -   C₂ is the N×(N−k) matrix whose columns are the N−k unused S-CDMA        codes.    -   v is a length N column vector representing added burst noise        samples, one sample per chip.

Here, we note that the despreader output corresponding to the unusedcodes contains the contribution of impulse noise samples, including theimpulse noise samples, and no signal contribution, i.e.,C*₂r=C*₂vFurthermore, we can write:

$y = {{C^{*}r} = {\begin{bmatrix}y_{1} \\y_{2}\end{bmatrix} = \begin{bmatrix}{x + {C_{1}^{*}v}} \\{C_{2}^{*}v}\end{bmatrix}}}$

Now, in order to cancel impulse noise, we need to estimate the termC*₁v. To do so, we use the noise-only term y₂=C*₂v. We now takeadvantage of the fact that for burst noise, only m samples of the noisevector v are of significant amplitude. The remaining N−m samples areassumed to be small enough that they can be ignored, and treated aszeros. This is a reasonable model for many cases of burst noise, whichit is assumed to be localized in its time support. We further assumethat we know which elements of the burst noise v are nonzero. Forterminology, we may refer to the known positions of the burst noise asthe significant noise samples, and the other (assumed zero) noisesamples as the null noise samples. We now further prune the matrices andvectors to remove the elements corresponding to the null elements of v.We are left with the reduced system of equations:y₂C*_(v)v_(nz)Where:

-   -   y₂ is a length N−k column vector representing despread signal        plus noise samples (soft decisions at despreader output).    -   v_(nz) is a length in column vector of non-zero noise samples        only, where m is strictly less than or equal to the number of        unused codes (N−k).

C_(v) is the (N−k)×m matrix whose columns are the N−k unused S-CDMAcodes, with the rows corresponding to the null elements of v removed,leaving only the m chips in each code which correspond to thesignificant elements of v.

Using standard matrix optimization techniques, we can derive the optimalestimate for v_(nz) given y₂, in the least-squares sense. The optimalestimate is given by{circumflex over (v)} _(nz) =[C _(v) C* _(v)]⁻¹ C _(v) y ₂

We have now solved for the in estimated noise samples {circumflex over(v)}_(v), and earlier we assumed knowledge of their positions in thelength-N noise vector v. We can therefore reconstruct the full-sizenoise vector by placing zeros in the null positions and the estimatednoise samples in the significant positions. The result is the length-Nestimated burst-noise vector {circumflex over (v)}.

Now that we have estimated the noise, we may subtract it from thereceived vector r. This step removes the estimated interference from thereceived signal, that is, performs the desired cancellation. We thendespread r with the matrix of used codes to get the despread symbols.The equation for these operations is:{circumflex over (x)} ₁ =C* ₁(r−{circumflex over (v)})

This form of the equation shows the interference being removed beforethe despreader.

Equivalently, the multiplication by the despreading matrix can bedistributed over the subtraction to give:

$\begin{matrix}{{\hat{x}}_{1} = {{C_{1}^{*}r} - {C_{1}^{*}\hat{v}}}} \\{= {y_{1} - {C_{1}^{*}\hat{v}}}}\end{matrix}$

This form of the equation shows the interference being removed after thedespreader. Unlimited further forms of the equation may be derivedwithout departing from the spirit and scope of the invention.

Another variant embodiment of the present invention may be performed byapplying the linear combiner at the chip level instead of the de-spreadsymbol level. In this approach, the 128 chips (again using the 128 codeembodiment example) used in the de-spreader for the desired code areadapted (for example, using the LMS method, a LS method, or anotheriterative solution) until they converge to the near-optimal adaptedde-spreading code. This adapted de-spreading code will be complex-valuedand will be a linear combination of all 128 de-spreading codes. In thisapproach there are 128 adaptive complex weights that need to be trained,so convergence is slower than for the baseline approach in which, forexample, only 8 weights need to be trained for the 8 unused codes.

Other methods for determining the combining weights may also be employedother than iterative solutions such as LMS and LS. For example,predetermined sets of combining weights may be stored for expectedoperational cases. For example, when burst noise has a duration of 16chips or less of a spreading sequence, sets of combining weights may bestored and retrieved as required. In such case, a first set of combiningweights would be retrieved and applied to cancel burst noise thatcorrupts chips 1-16, a second set of combining weights would beretrieved and applied for use on burst noise that arrives during chips2-17, a third set of combining weights would be retrieved and appliedfor burst noise that arrives during chips 3-18, . . . , and an Nth setof combining weights would be retrieved and applied for burst noise thatarrives during chips 113-128. This implementation obviates real-timecomputation of combining weights for the expected operational cases.

The general applicability of the present invention across a wide varietyof contexts is to be understood. The present invention is operable tocancel not only narrowband interference, but any interference thatoccupies a small number of dimensions in the signal space. The firstexample, given above, is narrowband interference. The example that isemphasized in the present disclosure is impulse or burst noise. However,other types of interference may be substantially eliminated according tothe present invention as well. A narrowband signal occupies a smallnumber of DFT bins (DFT: discrete Fourier transform—one example of anorthonormal expansion) bins, showing that it occupies a small number ofdimensions in signal space.

An extremely simple example is a CW (Continuous Wave) signal whosefrequency is an integer multiple of the de-spread symbol rate; this CWsignal occupies only a single bin in the DFT, or only one dimension inthe signal space, where each dimension is, in this case, one DFT bin.Another example is a short burst of noise (impulse or burst noise). Ashort burst signal occupies a small number of time samples, againshowing that it too occupies a small number of dimensions, where eachdimension is, in this case, a time sample. Other types of signals can beconstructed without limit that satisfies the property that they occupy asmall number of dimensions. All such signals can be canceled by thepresent technique.

Again, the DFT is just one example of an orthonormal expansion. A secondexample is the code matrix in DOCSIS 2.0 S-CDMA. Innumerable otherorthonormal transforms exist. If a signal occupies a small number ofdimensions in any orthonormal transform, the present invention'stechnique will help cancel it.

The present invention may also be implemented to use unused spatialdimensions to cancel interference according to the present invention.For example, we can generalize the technique to spatial dimensions aswell. One such interpretation of spatial dimensions is with respect toMIMO (multi-input multi-output) systems. We may designate certaintransmit antennas to transmit zeros at certain times. At the receiver,we may also utilize the samples from extra receive antennas to cancelinterference.

In any of the embodiments described herein, the present invention isoperable to store pre-computed weights using any number of variousstorage techniques, such as a look up table (LUT), memory, or some otherstorage technique. This may be beneficial in some cases where it may beimpractical to compute the weights fast enough, so we may want topre-compute some “canned” sets of weights. As an example, consider verywideband interference that occupies ½ the bandwidth of the desiredsignal. Assuming an S-CDMA system, we will need to have 64 unused codesout of 128 total codes. Each desired code now has 64 adaptive weightsthat need to be solved for. This implies a very large matrix to invert,which is very complex to implement in real time. However, we note thatin our favor, there are very few notches of this size (half thebandwidth) to go across the band. If, for example, we pre-compute theweights for several wideband notches and store the weight sets, then wecan simply select the notch that most closely matches the interferencewhen it occurs.

The present invention is also operable to support adjusting and trackingof pre-computed weights. As an extension to the above concept of storingpre-computed weights, we may store a single prototype set of weights,and modify the weights to move the notch around. For example, we canadjust the frequency of the notch without having to completelyre-compute the weights. We can also adjust the depth of the notch. Wecan weight and superimpose pre-computed notches to build up a morecomplex notch structure. We can build a tracking loop that automaticallyadjusts the frequency (or other parameter) of the notch as theinterference changes. Say there is narrowband interference, and we haveapplied a notch that cancels it. Now let the interference slew itsfrequency. We can implement a tracking loop that automatically slews thefrequency location of the notch to track the frequency of theinterference. There are many ways to implement such a tracking loop. Oneway is by taking an FFT of the interferences and tracking the energy inthe peak corresponding to the interference. Another way is to dither thelocation of the notch, and measure the power or SNR at the output. Wethen move the notch in the direction of increasing SNR or decreasinginterference or total power. Many other tracking methods can be devised.

The present invention may also be implemented to perform adjacentchannel interference (ACI) cancellation when performing interferencecancellation. For example, the canceller can also be used to cancel ACI.Consider a desired signal with signals present in the upper and loweradjacent channels. If the adjacent channel signals overlap slightly withthe desired-signal band, ACI results. ACI is in general colored and cantherefore be canceled using the present canceller technique.

In performing the processing, the present invention may employ a slidingwindow. The processing has thus far been described as being done on ablock basis, where a block is typically a symbol of 128 chips in theDOCSIS 2.0 implementation. However, the present invention is alsooperable when employing a sliding block approach as well. This may havethe greatest benefit when the code matrix is the identity matrix, whereno spreading occurs.

Code hopping is defined in the DOCSIS 2.0 spec as a process whereby thecode matrix is modified by a cyclic rotation of the rows of the entirecode matrix (except possibly the all −1's code) on each spreadinginterval. This causes the set of unused codes to be different on eachspreading interval, requiring the re-computation of the adaptive weightsin the canceller on each spreading interval. It would be better to hopover the set of codes excluding the unused codes, so that the unusedcodes will be the same on each spreading interval. This would obviatethe need to re-computation of the adaptive weights on each spreadinginterval, reducing processing complexity. Or, we can just turn off codehopping when the canceller is used. However, this latter approachremoves the benefits of code hopping, which include fairness (equalityof average performance) for all users.

In the case of the DOCSIS 2.0 S-CDMA code matrix, the rows, or basisfunctions, are nearly cyclic shifts of each other, with the exception ofcode 0. This property relates adjacent codes to shifts in time by onechip. This in turn relates the linear combiner used on the de-spreaderoutputs to a FIR filter in the time domain. This implies that theadaptive weights for the unused codes are similar to the weights thatare produced by the time domain interference cancellation structure usedfor TDMA. The latter weights can be computed efficiently by the Trenchmethod. Hence we might use the Trench method to compute an initial setof weights for the subspace-based canceller, and iterate the weights toa more exact solution using the LMS or similar tracking method.

In addition, a decision feedback equalizer (DFE)-like structure may alsobe performed, as follows. The canceller begins by using the de-spreaderoutputs of codes 1-8 (for example) in the linear combiner to estimatethe output of the code 9 de-spreader. A hard decision is made on thesymbol from code 9. Now that the symbol on code 9 is available, it canbe used to estimate the symbol on code 10. Once the symbol on code 10 isavailable, it can be used to estimate the symbol on code 11, and so on.A DFE-like structure can be constructed that can run in both directions.

As mentioned and described above, the subspace canceller can be appliedto an arbitrary code matrix (basis set). One such basis set is thediscrete Fourier transform (DFT). In this context one should alsomention the fast DFT methods collectively referred to as the fastFourier transform (FFT). The DFT is not well suited for narrowbandinterference cancellation in the receiver without modifications to thecancellation approach. This is because each basis function, the complextone e^(jnω) (or alternatively written as exp(jnω)), is designed to haveminimal support in the frequency domain. The basis functions cannotcancel a narrowband interferer without very large weights. Instead,locating the canceller predominantly in the transmitter is a betterchoice. One simply does not transmit those tones that overlap with theinterference, designating them as unused “codes” or tones. This approachhas been known in the industry for some time in conjunction withdiscrete multi-tone (DMT) or orthogonal frequency division multiplexing(OFDM) systems.

Moreover, the subspace canceller can be integrated with FEC coding. Inone approach, a Reed-Solomon code can provide parity symbols that aretransmitted on the unused spreading codes instead of zero symbols. Inanother approach, different SNRs may exist on different spreading codes(for example, when spreading code hopping is turned off). In this caseunequal transmitted power may be sent on each spreading code, or unequalnumbers of bits per symbol of modulation. Or, unequal coding strengthcan be used on each spreading code: for example, rate 7/8 on onespreading code and rate 3/4 on another spreading code. An importantpoint is that these approaches will give a better result than codehopping. In code hopping, performance is averaged over all spreadingcodes. It is better to use our prior knowledge of which spreading codesare disadvantaged, and give them more processing power or transmitpower.

The present invention is also operable, when performing interferencecancellation, to perform adjustment of transmitted spectrum to meet adesired spectral mask. It may be desirable to attenuate certain portionsof the spectrum using the subspace canceller. In these applications thecanceller is predominantly located in the transmitter. One example is ina wireless local area network (LAN) application. Here one has to place anotch in the transmit spectrum at the spectral location of the XMsatellite radio service. The subspace canceller has a great advantageover a notch filter. A notch filter can notch out a designated spectralregion, but in doing so, it distorts the desired signal. The subspacecanceller can create the notch without distorting the desired signal.The subspace canceller may also have the following advantage over anotch filter. A notch filter can cause large excursions in thetransmitted power, whereas the subspace canceller does not.

The interference cancellation according to the present invention mayalso be integrated with pre-coding. The subspace canceller can beintegrated with pre-coding, such as Tomlinson-Harashima pre-coding. Wenote that the subspace canceller can be used for narrowband interferencecancellation and also for equalization of a deep notch in the channel.These are two applications of pre-coding. Hence, if subspacecancellation and pre-coding were combined, we would get some of theadvantages of both.

In view of the above detailed description of the invention andassociated drawings, other modifications and variations will now becomeapparent. It should also be apparent that such other modifications andvariations may be effected without departing from the spirit and scopeof the invention.

1. An apparatus, comprising: a linear combiner that identifies anadapted rotated code based on burst noise affected chips within a chipsequence code of a spread signal; and a vector despreader that uses theadapted rotated code to despread a spread signal to produce a pluralityof data symbols that is substantially free of burst noise.
 2. Theapparatus of claim 1, wherein the linear combiner: scales a plurality ofnon-data-carrying codes using a plurality of adaptive weights, thatcorresponds to the burst noise affected chips within the chip sequencecode of a spread signal, thereby generating a plurality of scalednon-data-carrying codes; and combines a data-carrying code with theplurality of scaled non-data-carrying codes thereby generating theadapted rotated code.
 3. The apparatus of claim 2, further comprising: alook up table (LUT) that stores the plurality of adaptive weights; andwherein: the linear combiner retrieves the plurality of adaptive weightsfrom the LUT.
 4. The apparatus of claim 3, wherein: the plurality ofadaptive weights is a pre-computed plurality of adaptive weights; andthe pre-computed plurality of adaptive weights is one set of a pluralityof sets of pre-computed adaptive weights.
 5. The apparatus of claim 2,further comprising: a burst noise detector that processes the spreadsignal to identify the burst noise affected chips within the chipsequence code of the spread signal; and a weight computation module thatdetermines the plurality of adaptive weights based on the burst noiseaffected chips.
 6. The apparatus of claim 2, wherein: the plurality ofadaptive weights is a plurality of complex-valued weights.
 7. Theapparatus of claim 2, wherein: the linear combiner scales thedata-carrying code using at least one adaptive weight of the pluralityof adaptive weights before the data-carrying code is combined with theplurality of scaled non-data-carrying codes.
 8. The apparatus of claim2, wherein: the plurality of non-data-carrying codes is modulated usingnumerically zero-valued symbols.
 9. The apparatus of claim 1, wherein:the apparatus includes a transmitter and a receiver, that includes thelinear combiner and the vector despreader, to support bi-directionalcommunication via a communication channel to which the apparatuscouples; the receiver receives the spread signal from the communicationchannel; and the transmitter employs at least one additional code tospread at least one additional plurality of data symbols therebygenerating at least one additional spread signal that is transmitted viathe communication channel.
 10. The apparatus of claim 1, wherein: theapparatus is a wireless communication device.
 11. The apparatus of claim1, wherein: the spread signal is a code-division multiple access (CDMA)signal or a synchronous code-division multiple access (S-CDMA) signal.12. The apparatus of claim 1, wherein: the plurality of data symbols ismodulated according to at least one of Binary Phase Shift Keying (BPSK),Quadrature Phase Shift Keying (QPSK) or Quadrature Amplitude Modulation(QAM), 8 Phase Shift Keying (8-PSK), 16 QAM, 32 QAM, 64 QAM, 128 QAM,256 QAM, 516 QAM, and 1024 QAM.
 13. An apparatus, comprising: atransmitter module that employs a first code to spread a first pluralityof symbols thereby generating a first spread signal that is transmittedvia a communication channel; and a receiver module, that includes alinear combiner and a vector despreader, that receives a second spreadsignal from the communication channel; and wherein: the linear combiner:scales a plurality of non-data-carrying codes using a plurality ofadaptive weights, that corresponds to burst noise affected chips withina chip sequence code of the second spread signal, thereby generating aplurality of scaled non-data-carrying codes; and combines adata-carrying code with the plurality of scaled non-data-carrying codesthereby generating an adapted rotated code; and the vector despreaderuses the adapted rotated code to despread the second spread signal toproduce a second plurality of data symbols that is substantially free ofburst noise.
 14. The apparatus of claim 13, wherein the receiver modulesincludes: a burst noise detector that processes the second spread signalto identify the burst noise affected chips within the chip sequence codeof the second spread signal; and a weight computation module thatdetermines the plurality of adaptive weights based on the burst noiseaffected chips.
 15. The apparatus of claim 13, wherein: the plurality ofadaptive weights is a plurality of complex-valued weights.
 16. Theapparatus of claim 13, further comprising: a look up table (LUT) thatstores the plurality of adaptive weights; and wherein: the linearcombiner retrieves the plurality of adaptive weights from the LUT. 17.The apparatus of claim 16, wherein: the plurality of adaptive weights isa pre-computed plurality of adaptive weights; and the pre-computedplurality of adaptive weights is one set of a plurality of sets ofpre-computed adaptive weights.
 18. The apparatus of claim 13, wherein:the linear combiner scales the data-carrying code using at least oneadaptive weight of the plurality of adaptive weights before thedata-carrying code is combined with the plurality of scalednon-data-carrying codes.
 19. The apparatus of claim 13, wherein: theplurality of non-data-carrying codes is modulated using numericallyzero-valued symbols.
 20. The apparatus of claim 13, wherein: theapparatus is a wireless communication device.
 21. The apparatus of claim13, wherein: at least one of the first spread signal and the secondspread signal is a code-division multiple access (CDMA) signal or asynchronous code-division multiple access (S-CDMA) signal.
 22. Theapparatus of claim 13, wherein: the first plurality of data symbols andthe second plurality of data symbols is modulated according to at leastone of Binary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying(QPSK) or Quadrature Amplitude Modulation (QAM), 8 Phase Shift Keying(8-PSK), 16 QAM, 32 QAM, 64 QAM, 128 QAM, 256 QAM, 516 QAM, and 1024QAM.
 23. A method, comprising: receiving a spread signal, that hasincurred burst noise, from a communication channel; within a linearcombiner, identifying an adapted rotated code based on burst noiseaffected chips within a chip sequence code of the spread signal; andwithin a vector despreader, employing the adapted rotated code todespread a spread signal to produce a plurality of data symbols that issubstantially free of burst noise.
 24. The method of claim 23, furthercomprising: processing the spread signal to identify the burst noiseaffected chips within the chip sequence code of the second spreadsignal; determining a plurality of adaptive weights based on the burstnoise affected chips; scaling a plurality of non-data-carrying codesusing the plurality of adaptive weights thereby generating a pluralityof scaled non-data-carrying codes; and combining a data-carrying codewith the plurality of scaled non-data-carrying codes thereby generatingthe adapted rotated code.
 25. The method of claim 24, wherein: theplurality of adaptive weights is a plurality of complex-valued weights.26. The method of claim 24, further comprising: scaling thedata-carrying code using at least one adaptive weight of the pluralityof adaptive weights before the data-carrying code is combined with theplurality of scaled non-data-carrying codes.
 27. The method of claim 24,further comprising: retrieving the plurality of adaptive weights from alook up table (LUT).
 28. The method of claim 27, wherein: the pluralityof adaptive weights is a pre-computed plurality of adaptive weights; andthe pre-computed plurality of adaptive weights is one set of a pluralityof sets of pre-computed adaptive weights.
 29. The method of claim 24,wherein: the plurality of non-data-carrying codes is modulated usingnumerically zero-valued symbols.
 30. The method of claim 23, wherein:the spread signal is a code-division multiple access (CDMA) signal or asynchronous code-division multiple access (S-CDMA) signal.
 31. Themethod of claim 23, wherein: the plurality of data symbols is modulatedaccording to at least one of Binary Phase Shift Keying (BPSK),Quadrature Phase Shift Keying (QPSK) or Quadrature Amplitude Modulation(QAM), 8 Phase Shift Keying (8-PSK), 16 QAM, 32 QAM, 64 QAM, 128 QAM,256 QAM, 516 QAM, and 1024 QAM.
 32. An apparatus, comprising: a vectordespreader that employs a plurality of codes to despread a spreadsignal, that has incurred burst noise, thereby generating a plurality ofdespread signals; and a linear combiner that: scales a subset of theplurality of despread signals using a plurality of adaptive weightsthereby generating a plurality of scaled despread signals; and adds aselected one of the plurality of despread signals, which is not includedin the subset of the plurality of despread signals, to the plurality ofscaled despread signals thereby generating a modified despread signalfrom which a plurality of data symbols that is substantially free ofburst noise is produced.
 33. The apparatus of claim 32, wherein thelinear combiner: scales at least one additional subset of the pluralityof despread signals using at least one additional plurality of adaptiveweights thereby generating at least one additional plurality of scaleddespread signals; and adds at least one additional selected one of theplurality of despread signals, which is not included in the subset ofthe plurality of despread signals or the at least one additional subsetof the plurality of despread signals, to the plurality of scaleddespread signals and to the at least one additional plurality of scaleddespread signals thereby generating the modified despread signal. 34.The apparatus of claim 32, further comprising: a slicer that makes ahard decision corresponding to at least one of the plurality of datasymbols; and wherein: the linear combiner employs the hard decision as aknown symbol within at least one of the plurality of despread signalswhen generating at least one additional modified despread signal fromwhich at least one additional plurality of data symbols that issubstantially free of burst noise is produced.
 35. The apparatus ofclaim 34, wherein: the known symbol within the at least one of theplurality of despread signals is a non-zero or known non-data carryingsymbol.
 36. The apparatus of claim 34, wherein: the at least oneadditional plurality of data symbols includes less burst noise than theplurality of data symbols.
 37. The apparatus of claim 34, wherein: theplurality of data symbols is generated in a first iteration of iterativeprocessing; and the at least one additional plurality of data symbols isgenerated in a second iteration of the iterative processing.
 38. Theapparatus of claim 32, wherein: the subset of the plurality of despreadsignals corresponds to a plurality of non-data-carrying codes of theplurality of codes; and the selected one of the plurality of despreadsignals corresponds to a data-carrying code.
 39. The apparatus of claim38, wherein: each despread signal in a subset of the plurality ofdespread signals includes a corresponding known sequence of symbols. 40.The apparatus of claim 32, further comprising: a slicer that processesthe modified despread signal thereby generating the plurality of datasymbols.
 41. The apparatus of claim 32, further comprising: a burstnoise detector that processes the spread signal to identify burst noiseaffected chips within the chip sequence code of the spread signal; and aweight computation module that determines the plurality of adaptiveweights based on the burst noise affected chips.
 42. The apparatus ofclaim 32, wherein: at least one of the plurality of despread signalsincludes a known sequence of symbols.
 43. The apparatus of claim 42,wherein: the known sequence of symbols includes numerically zero-valuedsymbols.
 44. The apparatus of claim 32, further comprising: a look uptable (LUT) that stores the plurality of adaptive weights; and wherein:the linear combiner retrieves the plurality of adaptive weights from theLUT.
 45. The apparatus of claim 44, wherein: the plurality of adaptiveweights is a pre-computed plurality of adaptive weights; and thepre-computed plurality of adaptive weights is one set of a plurality ofsets of pre-computed adaptive weights.
 46. The apparatus of claim 32,wherein: the apparatus is a wireless communication device.
 47. Theapparatus of claim 32, wherein: the spread signal is a code-divisionmultiple access (CDMA) signal or a synchronous code-division multipleaccess (S-CDMA) signal.
 48. The apparatus of claim 32, wherein: theplurality of data symbols is modulated according to at least one ofBinary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying (QPSK)or Quadrature Amplitude Modulation (QAM), 8 Phase Shift Keying (8-PSK),16 QAM, 32 QAM, 64 QAM, 128 QAM, 256 QAM, 516 QAM, and 1024 QAM.
 49. Anapparatus, comprising: a linear combiner that identifies a first adaptedrotated code and a second adapted rotated code based on burst noiseaffected chips within a chip sequence code of a spread signal; and avector despreader that: uses the first adapted rotated code, thateffectuates a first notch at a first location within the burst noiseaffected chips, to despread a spread signal to produce a first pluralityof data symbols having first burst noise; and uses the second adaptedrotated code, that effectuates a second notch at a second locationwithin the burst noise affected chips, to despread the spread signal toproduce a second plurality of data symbols having second burst noise;and wherein: when the first burst noise is less than the second burstnoise, the apparatus outputs the first plurality of data symbols; andwhen the first burst noise is greater than or equal to the second burstnoise, the apparatus outputs the second plurality of data symbols. 50.The apparatus of claim 49, wherein: the vector despreader superimposesthe first notch at the first location and the second notch at the secondlocation to despread the spread signal to produce a third plurality ofdata symbols having third burst noise.
 51. The apparatus of claim 49,wherein: the first notch at the first location is a first pre-computednotch; and the second notch at the second location is a secondpre-computed notch.
 52. The apparatus of claim 49, wherein: the secondnotch at the second location is a dithered version of the first notch atthe first location.
 53. The apparatus of claim 49, wherein: the firstnotch at the first location has a first depth; and the second notch atthe first location has a second depth.
 54. The apparatus of claim 49,wherein the linear combiner: scales a plurality of non-data-carryingcodes using a plurality of adaptive weights, that corresponds to theburst noise affected chips within the chip sequence code of a spreadsignal, thereby generating a plurality of scaled non-data-carryingcodes; and combines a data-carrying code with the plurality of scalednon-data-carrying codes thereby generating at least one of the firstadapted rotated code and the second adapted rotated code.
 55. Theapparatus of claim 54, further comprising: a look up table (LUT) thatstores the plurality of adaptive weights; and wherein: the linearcombiner retrieves the plurality of adaptive weights from the LUT. 56.The apparatus of claim 55, wherein: the plurality of adaptive weights isa pre-computed plurality of adaptive weights; and the pre-computedplurality of adaptive weights is one set of a plurality of sets ofpre-computed adaptive weights.
 57. The apparatus of claim 49, wherein:the second adapted rotated code is identified based on at least onecharacteristic of the first plurality of data symbols having first burstnoise.
 58. The apparatus of claim 57, wherein: the at least onecharacteristic of the first plurality of data symbols having first burstnoise is a signal to noise ratio (SNR).
 59. The apparatus of claim 49,wherein: the apparatus is a wireless communication device.
 60. Theapparatus of claim 49, wherein: the spread signal is a code-divisionmultiple access (CDMA) signal or a synchronous code-division multipleaccess (S-CDMA) signal.
 61. The apparatus of claim 49, wherein: theplurality of data symbols is modulated according to at least one ofBinary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying (QPSK)or Quadrature Amplitude Modulation (QAM), 8 Phase Shift Keying (8-PSK),16 QAM, 32 QAM, 64 QAM, 128 QAM, 256 QAM, 516 QAM, and 1024 QAM.